Cyclic error compensation in interferometry systems

ABSTRACT

A first portion of a beam including a first frequency component is directed along a first path. A second portion of the beam is frequency shifted to generate a shifted beam that includes a second frequency component different from the first frequency component and one or more spurious frequency components different from the first frequency component. At least a portion of the shifted beam is directed along a second path different from the first path. An interference signal S(t) from interference between the beam portions directed along the different paths is measured. The signal S(t) is indicative of changes in an optical path difference n{tilde over (L)}(t) between the paths, where n is an average refractive index along the paths, {tilde over (L)}(t) is a total physical path difference between the paths, and t is time. An error signal is provided to reduce errors in an estimate of {tilde over (L)}(t) that are caused by at least one of the spurious frequency components of the shifted beam, the error signal being derived at least in part based on the signal S(t).

BACKGROUND

This invention relates to interferometers, e.g., displacement measuringand dispersion interferometers that measure displacements of ameasurement object such as a mask stage or a wafer stage in alithography scanner or stepper system, and also interferometers thatmonitor wavelength and determine intrinsic properties of gases.

Displacement measuring interferometers monitor changes in the positionof a measurement object relative to a reference object based on anoptical interference signal. The interferometer generates the opticalinterference signal by overlapping and interfering a measurement beamreflected from the measurement object with a reference beam reflectedfrom the reference object.

In many applications, the measurement and reference beams haveorthogonal polarizations and different frequencies. The differentfrequencies can be produced, for example, by laser Zeeman splitting, byacousto-optical modulation, or internal to the laser using birefringentelements or the like. Some frequency shifting techniques are based onphase modulation. For example, Serrodyne modulation can be applied to abeam using an electro-optic modulator to impose a ramp (or “sawtooth”)phase modulation corresponding to a desired frequency shift. Theorthogonal polarizations allow a polarizing beam splitter to direct themeasurement and reference beams to the measurement and referenceobjects, respectively, and combine the reflected measurement andreference beams to form overlapping exit measurement and referencebeams. The overlapping exit beams form an output beam that subsequentlypasses through a polarizer. The polarizer mixes polarizations of theexit measurement and reference beams to form a mixed beam. Components ofthe exit measurement and reference beams in the mixed beam interferewith one another so that the intensity of the mixed beam varies with therelative phase of the exit measurement and reference beams. A detectormeasures the time-dependent intensity of the mixed beam and generates anelectrical interference signal proportional to that intensity. Becausethe measurement and reference beams have different frequencies, theelectrical interference signal includes a “heterodyne” signal portionhaving a beat frequency equal to the difference between the frequenciesof the exit measurement and reference beams. If the lengths of themeasurement and reference paths are changing relative to one another,e.g., by translating a stage that includes the measurement object, themeasured beat frequency includes a Doppler shift equal to 2vnp/λ, wherev is the relative speed of the measurement and reference objects, λ isthe wavelength of the measurement and reference beams, n is therefractive index of the medium through which the light beams travel,e.g., air or vacuum, and p is the number of passes to the reference andmeasurement objects. Changes in the relative position of the measurementobject correspond to changes in the phase of the measured interferencesignal, with a 2π phase change substantially equal to a distance changeL_(RT) of λ/(np), where L_(RT) is a round-trip distance change, e.g.,the change in distance to and from a stage that includes the measurementobject.

Unfortunately, this equality is not always exact. Many interferometersinclude nonlinearities such as what are known as “cyclic errors.” Thecyclic errors can be expressed as contributions to the phase and/or theintensity of the measured interference signal and have a sinusoidaldependence on the change in optical path length pnL_(RT). For example, afirst order harmonic cyclic error in phase has a sinusoidal dependenceon (2πpnL_(RT))/λ and a second order harmonic cyclic error in phase hasa sinusoidal dependence on 2 (2πpnL_(RT))/λ Additional cyclic errors mayinclude higher order harmonic cyclic errors, negative order harmoniccyclic errors, and sub-harmonic cyclic errors.

Cyclic errors can be produced by “beam mixing,” in which a portion of aninput beam that nominally forms the reference beam propagates along themeasurement path and/or a portion of an input beam that nominally formsthe measurement beam propagates along the reference path. Such beammixing can be caused by ellipticity in the polarizations of the inputbeams and imperfections in the interferometer components, e.g.,imperfections in a polarizing beam splitter used to direct orthogonallypolarized input beams along respective reference and measurement paths.Because of beam mixing and the resulting cyclic errors, there is not astrictly linear relation between changes in the phase of the measuredinterference signal and the relative optical path length pnL between thereference and measurement paths. If not compensated, cyclic errorscaused by beam mixing can limit the accuracy of distance changesmeasured by an interferometer. Cyclic errors can also be produced byimperfections in transmissive surfaces that produce undesired multiplereflections within the interferometer and imperfections in componentssuch as retroreflectors and/or phase retardation plates that produceundesired ellipticities in beams in the interferometer. For a generalreference on some theoretical causes of cyclic error, see, for example,C. W. Wu and R. D. Deslattes, “Analytical modelling of the periodicnonlinearity in heterodyne interferometry,” Applied Optics, 37,6696-6700, 1998.

Another cause of errors, including cyclic errors, is imperfections inthe techniques used to impose the frequency shift between themeasurement and reference beams. For example, when Serrodyne modulationis used and the sawtooth phase modulation applied to one of the beamshas a non-zero fall time and/or an amplitude that is not an exactmultiple of 2π, spurious frequency components can result. For examplesof some of the effects of imperfect Serrodyne modulation, see Laskoskieet al., “Ti—LiNbO3 Waveguide Serrodyne Modulator with Ultrahigh SidebandSuppression for Fiber Optic Gyroscopes.,” Journal of LightwaveTechnology, Vol. 7, No. 4, 600-606, April 1989.

In dispersion measuring applications, optical path length measurementsare made at multiple wavelengths, e.g., 532 nm and 1064 nm, and are usedto measure dispersion of a gas in the measurement path of the distancemeasuring interferometer. The dispersion measurement can be used toconvert the optical path length measured by a distance measuringinterferometer into a physical length. Such a conversion can beimportant since changes in the measured optical path length can becaused by gas turbulence and/or by a change in the average density ofthe gas in the measurement arm even though the physical distance to themeasurement object is unchanged. In addition to the extrinsic dispersionmeasurement, the conversion of the optical path length to a physicallength requires knowledge of an intrinsic value of the gas. The factor Πis a suitable intrinsic value and is the reciprocal dispersive power ofthe gas for the wavelengths used in the dispersion interferometry. Thefactor Π can be measured separately or based on literature values.Cyclic errors in the interferometer also contribute to dispersionmeasurements and measurements of the factor Π. In addition, cyclicerrors can degrade interferometric measurements used to measure and/ormonitor the wavelength of a beam.

The interferometers described above are often crucial components ofscanner systems and stepper systems used in lithography to produceintegrated circuits on semiconductor wafers. Such lithography systemstypically include a translatable stage to support and fix the wafer,focusing optics used to direct a radiation beam onto the wafer, ascanner or stepper system for translating the stage relative to theexposure beam, and one or more interferometers. Each interferometerdirects a measurement beam to, and receives a reflected measurement beamfrom, a plane mirror attached to the stage. Each interferometerinterferes its reflected measurement beams with a correspondingreference beam, and collectively the interferometers accurately measurechanges in the position of the stage relative to the radiation beam. Theinterferometers enable the lithography system to precisely control whichregions of the wafer are exposed to the radiation beam.

In practice, the interferometry systems are used to measure the positionof the wafer stage along multiple measurement axes. For example,defining a Cartesian coordinate system in which the wafer stage lies inthe x-y plane, measurements are typically made of the x and y positionsof the stage as well as the angular orientation of the stage withrespect to the z axis, as the wafer stage is translated along the x-yplane. Furthermore, it may be desirable to also monitor tilts of thewafer stage out of the x-y plane. For example, accurate characterizationof such tilts may be necessary to calculate Abbe offset errors in the xand y positions. Thus, depending on the desired application, there maybe up to five degrees of freedom to be measured. Moreover, in someapplications, it is desirable to also monitor the position of the stagewith respect to the z-axis, resulting in a sixth degree of freedom.

SUMMARY

Among other aspects, the invention features electronic processingmethods that characterize and compensate cyclic errors ininterferometric data. Because cyclic errors are compensatedelectronically, the interferometry system that produces the data hasgreater tolerance to optical, mechanical, and electronic imperfectionsthat can cause cyclic errors, without sacrificing accuracy. Thecompensation techniques are especially useful for interferometric dataused to position microlithographic stage systems.

In part, the invention is based on the realization that prior values ofa main interferometric signal can be used to reduce the effect of cyclicerrors on an estimate of a length being measured by the interferometrysystem (e.g., a length indicating the position of a stage). Any of anumber of signal transformations such as a quadrature signal or aFourier transform can be derived from the prior values of the maininterferometric signal. These derived signal transformations can then beused to generate one or more error basis functions that represent one ormore cyclic error terms in the main interferometric signal. Appropriateamounts of each error basis function (e.g., as determined by acoefficient for each error basis function) form an error signal that issubtracted from a signal from which the measured length is obtained. Thecorresponding reduction in cyclic errors increases the accuracy of themeasured length. Using the error signal and the resulting compensatedlength estimate, the position of the measurement object can be monitoredand/or controlled in a variety of applications.

For example, the prior values of the main interferometric signal can beused to calculate an estimate for a quadrature signal for the maininterferometric signal. Algebraic combinations of such signals can yielderror basis functions in the form of sinusoidal functions whosetime-varying arguments correspond to particular cyclic error terms. Inembodiments in which the interferometer beams have a heterodynefrequency splitting, one may also calculate the quadrature signal of theheterodyne reference signal, and the error basis functions may bederived from algebraic combinations the main signal, the referencesignal, and the quadrature signals of the main and reference signals.

The error basis functions are used to isolate particular cyclic errorterms in the main signal and characterize coefficients representative ofeach cyclic error term (e.g., its amplitude and phase). For example,algebraic combinations of the error basis functions and the main signaland its quadrature signal can move a selected cyclic error term tozero-frequency, where low-pass filtering techniques (e.g., averaging)can be used to determine its amplitude and phase. Such coefficients arestored. Thereafter, a superposition of the error basis functionsweighted by the stored coefficients can be used to generate an errorsignal that can be subtracted from the main signal to reduce the cyclicerrors therein and improve its accuracy.

The technique is particularly useful when the Doppler shift is smallrelative to the heterodyne frequency because the frequency of eachcyclic error term is nearly equal to that of primary component of themain signal, in which case the estimate for the quadrature signal of themain signal is more accurate. This is an especially important propertybecause it is precisely when the frequencies of the cyclic error termsare near that of the primary component of the main signal that thecyclic error terms are most problematic because they cannot be removedby frequency filtering techniques. Furthermore, at small Doppler shifts,one or more of the cyclic error frequencies may be within the bandwidthof a servo system used to position a stage based on the interferometricsignal, in which the case the servo loop may actually amplify the cyclicerror term when positioning the stage. Small Doppler shifts are actuallyquite common in microlithographic stage systems, such as when searchingfor an alignment mark, scanning in an orthogonal dimension to the onemonitored by the interferometric signal, and changing stage direction.Moreover, at small Doppler shifts, selecting an integral relationshipbetween the sampling rate of the detector and the heterodyne frequency(e.g., 6:1) yields an especially simple formula for the quadraturesignal.

In addition, at small Doppler shifts, the main signal is nearly periodicwith the heterodyne frequency, in which case prior data can be used togenerate the error signal. As a result, correction of the main signalcan be accomplished with only a single real-time subtraction of theerror signal from the main signal, significantly reducing thecomputation time associated with the correction and thereby reducingdata age errors in any servo system for position a microlithographystage.

Error basis functions can be derived from other combinations of priorvalues of the main interferometric signal besides a quadrature signalincluding a Fourier transform of the main interferometric signal. In thecase of a Fourier transform, the resulting error signal is subtractedfrom a complex signal from which the measured length is obtained, asdescribed in more detail below. In some cases (e.g., when the stage ismoving slowly or is nearly stationary), the error basis functions can bederived from a distribution of values, where each of the values isgenerated from multiple values of the main interferometric signal, andthe values in the distribution (e.g., multi-dimensional valuesdistributed over a multi-dimensional space such as a complex space withreal and imaginary dimensions) are not necessarily sequential in time.

We now summarize various aspects and features of the invention.

In one aspect, in general, the invention features a method that includesdirecting a first portion of a beam including a first frequencycomponent along a first path. The method includes frequency shifting asecond portion of the beam to generate a shifted beam that includes asecond frequency component different from the first frequency componentand one or more spurious frequency components different from the firstfrequency component. The method includes directing at least a portion ofthe shifted beam along a second path different from the first path. Themethod includes measuring an interference signal S(t) from interferencebetween the beam portions directed along the different paths. The signalS(t) is indicative of changes in an optical path difference n{tilde over(L)}(t) between the paths, where n is an average refractive index alongthe paths, {tilde over (L)}(t) is a total physical path differencebetween the paths, and t is time. The method includes providing an errorsignal to reduce errors in an estimate of {tilde over (L)}(t) that arecaused by at least one of the spurious frequency components of theshifted beam, the error signal being derived at least in part based onthe signal S(t).

Aspects of the invention can include one or more of the followingfeatures.

The method further comprises providing the estimate of {tilde over(L)}(t) based at least in part on subtracting the error signal from thesignal S(t).

Frequency shifting the second beam portion comprises modulating thephase of the second beam portion to generate the shifted beam.

The phase modulation comprises electro-optic phase modulation.

The phase modulation comprises Serrodyne modulation.

The errors are dependent on a non-zero fall time of a ramp associatedwith the Serrodyne modulation.

The errors are further dependent on an amplitude of the ramp.

The method further comprises providing the estimate of {tilde over(L)}(t) based at least in part on adjusting the phase modulation of thesecond beam portion using the error signal.

The phase modulation is adjusted using the phase of the error signal.

An amplitude of the phase modulation is adjusted using the phase of theerror signal.

At least one of the beam portions is directed to reflect from a movablemeasurement object before producing the interference signal S(t) togenerate a Doppler shift indicative of movement of the measurementobject.

The method further comprises providing the estimate of {tilde over(L)}(t) based at least in part on a sinusoidal component of the signalS(t) that has a time-varying argument corresponding to a sum of a termproportional to a difference between peak frequencies of the firstfrequency component and the second frequency component and a termproportional to a Doppler shift frequency.

The error signal includes a sinusoidal component that has a time-varyingargument corresponding to a difference between the term proportional tothe difference between the peak frequencies of the first frequencycomponent and the second frequency component and the term proportionalto the Doppler shift frequency.

The difference between the peak frequencies of the first frequencycomponent and the second frequency component is less than about 500 kHz.

The Doppler shift frequency is less than about 50 kHz.

One of the paths is associated with a position of a reference object andthe other path is associated with a position of a moveable measurementobject.

The position of the moveable object is controlled by a servo system.

The servo system controls the position of the measurement object basedon the signal S(t) and the error signal.

Providing the error signal to reduce the errors caused by at least onespurious frequency component in the shifted beam comprises providing oneor more coefficients representative of one or more errors that cause thesignal S(t) to deviate from an ideal expression of the form A₁cos(ω_(R)t+φ(t)+ζ₁), where A₁ and ζ₁ are constants, ω_(R) is an angularfrequency difference between the first frequency component and thesecond frequency component, and φ(t)=nk{tilde over (L)}(t), with k=2π/λand λ equal to a wavelength for the beam.

The deviation can be expressed as

${\sum\limits_{m,p}{A_{m,p}{\cos \left( {{\omega_{R}t} + {\frac{m}{p}{\phi (t)}} + \zeta_{m,p}} \right)}}},$

where p=1, 2, 3 . . . , and m is any integer not equal to p, and wherethe provided coefficients comprise information corresponding to at leastsome of A_(m,p) and ζ_(m,p).

The coefficients are derived at least in part based on a plurality ofsamples of the signal S(t).

The error signal is generated from the coefficients and one or moreerror basis functions derived at least in part from a plurality ofsamples of the signal S(t).

Each of the error basis functions is derived at least in part from alinear combination of samples of the signal S(t).

Each of the error basis functions corresponds to a function thatincludes one or more leading sinusoidal terms having a time-varyingargument that corresponds to a time-varying argument of an error termthat represents a portion of the deviation of S(t) from the idealexpression.

Reducing errors in the estimate of {tilde over (L)}(t) comprisesderiving the estimate of {tilde over (L)}(t) from a difference betweenthe error signal and a discrete Fourier transform of samples of S(t).

A lithography method for use in fabricating integrated circuits on awafer includes supporting the wafer on a moveable stage; imagingspatially patterned radiation onto the wafer; adjusting the position ofthe stage; and monitoring the position of the stage using aninterferometry system, wherein monitoring the position of the stagecomprises reducing errors in an estimate of a physical path differenceassociated with a position of a measurement object associated with thestage using the method described above.

A method for fabricating integrated circuits includes applying a resistto a wafer; forming a pattern of a mask in the resist by exposing thewafer to radiation using the lithography method described above; andproducing an integrated circuit from the wafer.

A lithography method for use in the fabrication of integrated circuitsincludes directing input radiation through a mask to produce spatiallypatterned radiation; positioning the mask relative to the inputradiation; monitoring the position of the mask relative to the inputradiation using an interferometry system, wherein monitoring theposition of the mask comprises reducing errors in an estimate of aphysical path difference associated with the position of the mask usingthe method described above; and imaging the spatially patternedradiation onto a wafer.

A method for fabricating integrated circuits includes applying a resistto a wafer; forming a pattern of a mask in the resist by exposing thewafer to radiation using the lithography method described above; andproducing an integrated circuit from the wafer.

A lithography method for fabricating integrated circuits on a waferincludes positioning a first component of a lithography system relativeto a second component of a lithography system to expose the wafer tospatially patterned radiation; and monitoring the position of the firstcomponent relative to the second component using an interferometrysystem. Monitoring the position of the first component comprisesreducing errors in an estimate of a physical path difference associatedwith a position of a measurement object associated with the firstcomponent using the method described above.

The first component comprises projection optics, and the secondcomponent comprises a stage for supporting the wafer.

A method for fabricating integrated circuits includes applying a resistto a wafer; forming a pattern of a mask in the resist by exposing thewafer to radiation using the lithography method described above; andproducing an integrated circuit from the wafer.

A method for fabricating a lithography mask includes directing a writebeam to a substrate to pattern the substrate; positioning the substraterelative to the write beam; and monitoring the position of the substraterelative to the write beam using an interferometry system. Monitoringthe position of the substrate comprises reducing errors in an estimateof a physical path difference associated with a position of ameasurement object associated with the substrate using the methoddescribed above.

An apparatus comprising a computer readable medium which duringoperation causes a processor to perform the method described above.

In another aspect, in general, the invention features an apparatus thatincludes an interometry system. During operation, the interometry systemdirects a first portion of a beam including a first frequency componentalong a first path; frequency shifts a second portion of the beam togenerate a shifted beam that includes a second frequency componentdifferent from the first frequency component and one or more spuriousfrequency components different from the first frequency component; anddirects at least a portion of the shifted beam along a second pathdifferent from the first path. The apparatus includes a detector thatmeasures an interference signal S(t) from interference between the beamportions directed along the different paths, wherein the signal S(t) isindicative of changes in an optical path difference n{tilde over (L)}(t)between the paths, where n is an average refractive index along thepaths, {tilde over (L)}(t) is a total physical path difference betweenthe paths, and t is time. The apparatus includes an electronicprocessor, which during operation receives the interference signal S(t)from the detector and provides an error signal to reduce errors in anestimate of {tilde over (L)}(t) that are caused by at least one of thespurious frequency components of the shifted beam, the error signalbeing derived at least in part based on the signal S(t).

Aspects of the invention can include one or more of the followingfeatures.

The second beam portion is frequency shifted in a phase modulator thatmodulates the phase of the second beam portion to generate the shiftedbeam.

The phase modulator comprises an electro-optic phase modulator.

A lithography system for use in fabricating integrated circuits on awafer includes a stage for supporting the wafer; an illumination systemfor imaging spatially patterned radiation onto the wafer; a positioningsystem for adjusting the position of the stage relative to the imagedradiation; and the apparatus described above for monitoring the positionof the wafer relative to the imaged radiation.

A method for fabricating integrated circuits includes applying a resistto a wafer; forming a pattern of a mask in the resist by exposing thewafer to radiation using the lithography system described above; andproducing an integrated circuit from the wafer.

A lithography system for use in fabricating integrated circuits on awafer includes a stage for supporting the wafer; and an illuminationsystem including a radiation source, a mask, a positioning system, alens assembly, and the apparatus described above. During operation thesource directs radiation through the mask to produce spatially patternedradiation, the positioning system adjusts the position of the maskrelative to the radiation from the source, the lens assembly images thespatially patterned radiation onto the wafer, and the apparatus monitorsthe position of the mask relative to the radiation from the source.

A method for fabricating integrated circuits includes applying a resistto a wafer; forming a pattern of a mask in the resist by exposing thewafer to radiation using the lithography system described above; andproducing an integrated circuit from the wafer.

A beam writing system for use in fabricating a lithography mask includesa source providing a write beam to pattern a substrate; a stagesupporting the substrate; a beam directing assembly for delivering thewrite beam to the substrate; a positioning system for positioning thestage and beam directing assembly relative one another; and theapparatus described above for monitoring the position of the stagerelative to the beam directing assembly.

A method for fabricating a lithography mask includes directing a beam toa substrate using the beam writing system described above; varying theintensity or the position of the beam at the substrate to form a patternin the substrate; and forming the lithography mask from the patternedsubstrate.

As used herein “algebraic combinations” means combinations of operands(e.g., real or complex numbers including values of signals) according toone or more algebraic operations (e.g., addition, subtraction,multiplication, and division).

Unless otherwise defined, all technical and scientific terms used hereinhave the same meaning as commonly understood by one of ordinary skill inthe art to which this invention belongs. In case of conflict withpublications, patent applications, patents, and other referencesmentioned incorporated herein by reference, the present specification,including definitions, will control.

Other features and advantages of the invention will be apparent from thefollowing detailed description.

DESCRIPTION OF DRAWINGS

FIG. 1 is a block diagram of an interferometric position measuringsystem.

FIGS. 2 a and 2 b are plots of a phase modulation and a resultinginterference signal, respectively.

FIGS. 3 a-3 d are plots of a spectrum of an interference signal.

FIG. 4 a is a schematic diagram of a processing unit for generatingcyclic error basis functions and characterizing cyclic errorcoefficients based on a main interference signal S(t) and a referencesignal S_(R)(t).

FIG. 4 b is a schematic diagram of a processing unit for generating anerror signal S_(ψ)(t) from the cyclic error basis functions andcharacterized coefficients and using the error signal to reduce cyclicerrors in the main interference signal S(t).

FIG. 4 c is a schematic diagram of an exemplary measurement system forgenerating cyclic error basis functions and characterizing cyclic errorcoefficients based on a complex measurement signal.

FIG. 4 d is a schematic diagram of an error estimator for themeasurement system of FIG. 4 c.

FIG. 4 e is a schematic diagram of a processing unit for the errorestimator of FIG. 4 d.

FIG. 5 is a schematic diagram of an M^(th) order digital filter for usein low-pass filtering algebraic combinations of the main signal, thereference signal, their quadrature signals, and the error basisfunctions to yield the cyclic error coefficients.

FIG. 6 is a schematic diagram of an interferometry system including ahigh-stability plane mirror interferometer (HSPMI).

FIG. 7 is a schematic diagram of an embodiment of a lithography toolthat includes an interferometer.

FIG. 8 a and FIG. 8 b are flow charts that describe steps for makingintegrated circuits.

FIG. 9 is a schematic of a beam writing system that includes aninterferometry system.

DETAILED DESCRIPTION

Referring to FIG. 1, an exemplary interferometric position measurementsystem 1 includes a source 2 (e.g., a laser) that provides a first inputbeam whose spectrum has a frequency component with a narrow linewidthand a peak frequency corresponding to the optical wavelength of the beam(e.g., 1550 nm). A frequency shifter 4 frequency shifts a portion of thefirst input beam to generate a second frequency shifted input beam thatincludes a frequency component whose peak frequency is shifted relativeto the peak frequency of the first input beam. The system 1 alsoincludes an interferometer 6 to provide an interference signal thatrepresents an interferometric measurement of the position of ameasurement object, and a detection system 8 to detect and analyze theinterference signal to obtain an estimate of the position of themeasurement object.

The frequency shifter 4 can use any of a variety of techniques togenerate the measurement and reference beams depending, for example, onthe magnitude of the desired frequency difference between the beams. Insome systems, the beat frequency is on the order of a few MHz or tens ofMHz. In some systems, e.g. when the maximum Doppler frequency is low, alower beat frequency may be desirable. Circuitry operating at lowerfrequencies may have advantages of cost, noise, or dynamic range. Forexample, in a system with a maximum Doppler frequency of 20 kHz, a beatfrequency on the order of 100 kHz might be preferred. One approach forgenerating measurement and reference beams that differ by a low beatfrequency uses an electro-optic modulator (EOM) for Serrodynemodulation, which applies a sawtooth phase modulation to an input beamto generate the modulated beam. One advantage of using an EOM in suchsystems is that there are readily available low cost and reliabledevices (e.g., optical telecommunications devices) that include usefulfeatures such as low-insertion loss optical fiber couplers. In additionto cases in which a low beat and/or Doppler frequency is being used,Serrodyne modulation may also be desirable for other reasons for bothlow and high frequencies (e.g., beat frequency of a few or tens of MHzor more), for example, ruggedness, reliability, wavelength, solid statelaser, low cost of EOMs, etc.

The interferometer 6 accepts the first and second input beams anddirects them as measurement and reference beams along measurement andreference paths of the interferometer. Either of the input beams canserve as the measurement beam or the reference beam. The interferometer6 includes the measurement object whose position is being measuredrelative to a reference location defined by the interferometer 6. Forexample, the measurement object can include a mirror from which themeasurement beam is reflected at some point along the measurement path.

The detection system 8 detects optical interference between the twobeams in the interferometer 6 to generate an electronic interferencesignal S(t) that can be processed electronically, for example, afterdigital sampling of the signal. The signal S(t) is indicative of changesin an optical path difference n{tilde over (L)}(t) between the paths,where n is an average refractive index along the paths, L(t) is a totalphysical path difference between the paths, and t is time. The detectionsystem 8 is able to reduce errors in an estimate of {tilde over (L)}(t)that result from various causes using appropriate techniques. As anexample, we will consider errors due to imperfect Serrodyne modulationin the frequency shifter 4 as a cause of errors.

The interference signal S(t) can be expressed as a main sinusoidal termrepresenting a Doppler shifted measurement component plus an additionalerror term representing one or more error components as:

S(t)=A ₁ cos(φ_(R)+φ)+S _(ψ)(t).

The time dependent phase φ_(R) is a reference phase resulting from thefrequency difference (or “beat frequency”) between the measurement beamand the reference beam, with the beat frequency f_(R) given by|dφ_(R)/dt|=ω_(R)=2πf_(R). The time dependent phase φ represents theDoppler shift due to movement of a measurement object that is part ofthe interferometer 6, with the Doppler frequency f_(D) given by|dφ/dt|=ω_(D)=2πf_(D). S_(ψ)(t) is an error term that represents errorcomponents including cyclic error components and other error componentssuch as components at or near the harmonic frequencies of the beatfrequency.

Referring to FIG. 2 a, the frequency shifter 4 modulates the opticalpath length (e.g., by electro-optically modulating the index ofrefraction) to generate a periodic sawtooth phase pattern 100 that has asubstantially linear rise and a fast fall over a period T. The amplitudeA of the imposed phase shift (in units of radians) is selected to besubstantially equal to a multiple of 2π (e.g., within a few percent)A≈2πm. In some cases, a small multiple is selected (e.g., m=1) becausethe frequency shifter 4 may have a maximum voltage or electrical powerrating, or the frequency shifter 4 may have increased nonlinearities atthe larger voltage range needed for a larger phase shift, for example.The imposed sawtooth phase modulation results in beat frequency that isthen given by m/T. Thus, the beat frequency can be tuned by changing theinteger m and/or the period T.

FIG. 2B shows an interference pattern 101 over a period T of anexemplary interference signal S(t) detected from the interferometer 6driven by an input beam that has been modulated with the phase pattern100. The slower sinusoidal shape 102 during the rise of the phasepattern 100 corresponds to the desired beat frequency plus a Dopplerfrequency from movement of the measurement object. The faster sinusoidalshape 103 during the fall of the phase pattern 100 corresponds to“unwinding” over the short fall time of the phase accumulated during therise, plus the same Doppler shift. This high frequency “flyback”generates spurious frequency components in the spectrum of theinterference signal S(t) that appear at multiple frequencies.

FIG. 3A shows a spectrum 80 that corresponds to a sawtooth phasemodulation with a zero fall time and an amplitude of A=2π. In thisexample, the beat frequency is

$f_{R} = {\frac{1}{T} = {100\mspace{14mu} {kHz}}}$

and the Doppler frequency is f_(D)=10 kHz. There is a measurementfrequency component 82 at the Doppler shifted measurement frequency off_(R)+f_(D)=110 kHz.

FIG. 3B shows a spectrum 84 that corresponds to a sawtooth phasemodulation with a non-zero fall time of 200 ns and an amplitude of A=2π.The beat frequency and Doppler frequency are the same as in the previousexample of FIG. 3A (and in the examples of FIGS. 3C and 3D). In thisexample, in addition to a Doppler shifted measurement frequencycomponent 86, there are additional “spurious” frequency components duemainly to the flyback portions of the phase modulation. Otherimperfections that can affect the spurious frequency components include,for example, EOM drive amplitude value, EOM drive amplifier nonlinearityand EOM nonlinearity. Since the flybacks occur periodically, repeatingat time intervals of T, the peak frequencies of these spuriouscomponents appear near the beat frequency

$f_{R} = \frac{1}{T}$

and near harmonics of this frequency 2/T, 3/T, 4/T, etc. The peakfrequencies also depend on the Doppler shift which determines the actualphase accumulated in the sawtooth period. Thus, a “negative Doppler”frequency component 88 occurs at a frequency of f_(R)−f_(D)=90 kHz, andpairs of frequency components also occur at frequencies

${\frac{n}{T} \pm f_{D}},$

where n=2, 3, 4, etc. A more detailed derivation of similar frequencycharacteristics is available, for example, in Voges et al., “Opticalphase and amplitude measurement by single sideband homodyne detection”IEEE Journal of Quantum Electronics, Vol. 18, pages 124-129, 1982,incorporated herein by reference.

Most of these spurious frequency components can be compensated for usingbandpass filtering techniques. For example, the closest of these otherfrequency components appears at a frequency of

${\frac{2}{T} - f_{D}} = {190\mspace{14mu} {{kHz}.}}$

Even at small Doppler shifts (relative to the beat frequency) due to lowspeed measurement object movement, there remains a distance of almostf_(R)=100 kHz between the measurement frequency component 86 and thisspurious component. However, at small Doppler shifts, the negativeDoppler frequency component 88 could be so close to the measurementfrequency component 86 that filtering techniques could be ineffective orimpractical.

In such cases, the negative Doppler frequency component 88 can be moreeasily compensated for using the cyclic error compensation techniquesdescribed herein. Cyclic errors appear at frequencies that are shiftedfrom the beat frequency by multiples of the Doppler frequency, and canbe classified according to the frequency offset from the reference phaseφ_(R) as modeled in Equations (3)-(6) below. The negative Dopplerfrequency component 88 corresponds to a cyclic error of order “−1” withtime dependent phase φ_(R)−φ (modeled in Equation (3)). Other cyclicerror orders, which are not significant in this example, include order“0” with time dependent phase φ_(R) (modeled in Equation (4)), order “2”with time dependent phase φ_(R)+2φ (modeled in Equation (5)), and order“3” with time dependent phase φ_(R)+3φ (modeled in Equation (6)). Thus,while the techniques described herein for compensating for cyclic errorsare capable of compensating for multiple orders of cyclic errors, inthis example the single cyclic error of order “−1” is corrected usingthese techniques. In some systems, there may also be cyclic errors ofvarious orders due to other causes (e.g., optical) that maysimultaneously be compensated for by these techniques. By forming anerror signal that can be subtracted from the measurement signal, moreaccurate compensation of the errors affecting the estimate of {tildeover (L)}(t) can be achieved in some cases than can be achieved throughtechniques like bandpass filtering.

The magnitude of the spurious frequency components can also be reducedby adjusting characteristics of the phase modulation. In addition toreducing the fall time of the sawtooth, the magnitude of the spuriousfrequency components can be reduced by adjusting the amplitude of thesawtooth A away from an exact 2π phase shift by small amounts (e.g., bya few percent). FIG. 3C shows a spectrum 90 that corresponds to asawtooth phase modulation with a non-zero fall time of 200 ns and anamplitude of A=2π(1.01). With this increase of 1% in the amplitude A,the magnitude of the largest spurious component 92 is reduced relativeto the magnitude of the measurement component 91. FIG. 3D shows aspectrum 94 that corresponds to a sawtooth phase modulation with anon-zero fall time of 200 ns and an amplitude of A=2π(1.04). With thisincrease of 4% in the amplitude A, the magnitude of the negative Dopplerfrequency component 96 is reduced relative to the magnitude of themeasurement component 95. In this case, it is evident that adjusting theamplitude A to minimize the negative Doppler frequency component 96comes at the expense of increasing the amplitudes of some of thecomponents at the higher harmonics. However, since the effects of thecomponents at the higher harmonics can be reduced using filteringtechniques, it can be useful to minimize the negative Doppler component.Other adjustments to the Serrodyne phase modulation can be made toattempt to further reduce the negative Doppler component perhaps at theexpense of other error components, such as adjusting the shape of theramp of the sawtooth away from perfect linearity, for example.

The use of CEC enables measuring both the magnitude and phase of theerror components. The phase, which is related to amplitude A,facilitates direct adjustment of the modulation magnitude to minimizethe error magnitudes. The phase of the negative Doppler component may bedetermined by ζ⁻¹=arg(A⁻¹,B⁻¹) using the values in equations 37 and 38,or equivalently by ζ⁻¹=arg(C_(4R)) using the value in equation 102. Thisphase will exhibit a 2π phase change as the ramp gain is adjusted frombelow the optimum value to above the optimum value. Thus, the phase maybe used as an indicator of whether to increase or decrease the rampgain.

Embodiments include an electronic cyclic error compensation (CEC)procedure for compensation of cyclic error effects in interferometryapplications, such as heterodyne interferometry. In preferredembodiments, the compensation is achieved for low slew rates of a planemirror measurement object attached to a stage or attached to a referencesystem with associated interferometers attached to a stage. When opticaltechniques are used to eliminate and/or reduce the amplitudes of certaincyclic errors such as sub-harmonic cyclic errors to ≲0.05 nm (3σ), theremaining cyclic errors of the harmonic type with amplitudes of 0.5 nmor less can be treated as having constant amplitudes with fixed offsetphases and the required accuracy of the cyclic error compensation for aremaining cyclic error term is approximately 10% to meet a compensatedcyclic error budget of 0.05 nm (3σ) or less. Further, the number ofcyclic error terms that need to be compensated electronically aretypically a small number, e.g., of the order of 3. In preferredembodiments, the processing operations of CEC at high digital processingrates can be limited to a single add operation, whereas the remainingprocessing operations, which require additions, subtractions,multiplications, and divisions, can be performed at lower rates usingprior values of the interference signal.

Typically, cyclic error effects in heterodyne interferometry can beeliminated by filtering the heterodyne signal in frequency space, e.g.,using Fourier spectral analysis, when the Doppler shift frequencies canbe resolved by the frequency resolution of a phase meter used todetermine the heterodyne phase. Unfortunately, such filtering techniquescannot be used to eliminate cyclic error effects at low slew rates ofthe stage (including, e.g., zero speed of the stage) when thecorresponding Doppler shift frequencies cannot be distinguished from thefrequency of the primary signal. Further complications arise with cyclicerror frequencies are within the bandwidth of the servo system, in whichcase the cyclic errors can be coupled directly into the stage positionthrough the servo control system, and even amplify the error in thestage position from a desired position.

Specific details of embodiments of the CEC are described further below.In one approach, the CEC procedure processes real time-sampled values ofa digitized measurement signal (DMS) generated by ananalog-to-digital-converter (ADC). Advantages of this “DMS approach”include a cyclic error correction signal that may be generated in a“feed forward mode,” where the feed forward mode can involve a simplediscrete transform based on a translation in time and need not require aspectral analysis or the use of a discrete transform such as a discreteFourier transform, such as a fast Fourier transform (FFT). Likewise,conjugated quadratures of the main interference signal and the referencesignal can be generated by simple discrete transforms and need notrequire the use of a discrete transform such as a discrete Hilberttransform. Moreover, the feed forward mode can reduce the number ofcomputer logic operations that are required at the very highest computerates and can thereby reduce errors in data age that are introduced byincorporation of CEC.

In another approach, the CEC procedure processes complex values of acomplex measurement signal (CMS) generated by a discrete Fouriertransform (DFT) module after the ADC module. Advantages of this “CMSapproach” include the ability to update the DFT (and the CECcomputations) at a lower rate (e.g., 10 MHz) than the ADC sampling rate(e.g., 120 MHz). A reduction in the CEC update rate enables a simplifiedhardware architecture. For example, a reduction in CEC update rate by afactor of 12 can result in a hardware savings of greater than a factorof 12. The CMS approach also eliminates cyclic errors that are due tofinite arithmetic precision of the samples generated by the ADC and ofthe DFT coefficients and calculations in the DFT module. The CMSapproach is also less subject to noise than the DMS approach due to thenumber of samples and the window function used by the DFT module.

Another advantage of both the DMS approach and the CMS approach is thatthe cyclic error coefficients can be characterized at Doppler shiftfrequencies for which the phase meter cannot distinguish between thecyclic error frequencies from the frequency of the primary component ofthe interference signal. Furthermore, the cyclic error coefficients canbe characterized and used for compensation over a range of Doppler shiftfrequencies that is small relative to heterodyne frequency, which is arange over which the cyclic error coefficients are typically frequencyindependent, thereby simplifying the cyclic error correction.

We now describe the DMS approach for the CEC that operates in a feedforward mode of operation, in which a cyclic error correction signalS_(ψ)(t) is subtracted from a corresponding electrical interferencesignal S(t) of an interferometer to produce a compensated electricalinterference signal. The phase of the compensated electricalinterference signal is then measured by a phase meter to extractrelative path length information associated with the particularinterferometer arrangement. Because cyclic error effects have beenreduced, the relative path length information is more accurate. As aresult, the compensated electrical interference phase can be used tomeasure and control through a servo control system the position of astage, even at low slew rates including a zero slew rate where cyclicerror effects can otherwise be especially problematic. The DMS approachis also described in published U.S. application Ser. No. 10/616,504(publication number US 2004/0085545 A1).

Referring to FIGS. 4 a and 4 b, in a preferred embodiment, the CECcomprises two processing units. One processing unit 10 determines cyclicerror basis functions and factors relating to the amplitudes and offsetphases of cyclic errors that need be compensated. A second processingunit 60 of CEC generates cyclic error correction signal S_(ψ)(t) usingthe cyclic error basis functions and the factors relating to theamplitudes and offset phases determined by first processing unit 10. Thefirst processing unit 10 of CEC for the first embodiment is shownschematically in FIG. 4 a and the second processing unit 60 of CEC ofthe first embodiment is shown schematically in FIG. 4 b.

Referring now to FIG. 4 a, an optical signal 11 from an interferometeris detected by detector 12 to generate an electrical interferencesignal. The electrical interference signal is converted to digitalformat by an analog to digital converter (ADC) in converter/filter 52 aselectrical interference signal S(t) and sent to the CEC processor. Forexample, the ADC conversion rate is a high rate, e.g., 120 MHz.

In the present embodiment, we focus on a particular set of four cyclicerror terms that are compensated at low slew rates. Adaptation tocompensate for a different set of cyclic errors will be evident to oneskilled in the art based on the subsequent description. The electricalinterference signal S(t) comprising the four cyclic error terms can beexpressed in the form

S(t)=A ₁ cos(φ_(R)+φ+ζ₁)+S _(ψ)(t)  (1)

where

S _(ψ)(t)=S _(ψ−1)(t)+S _(ψ0) +S _(ψ2)(t)+S _(ψ3)(t);  (2)

S _(ψ−1)(t)=ε⁻¹ cos(φ_(R)−φ+Λ⁻¹),  (3)

S _(ψ0)=ε₀ cos(φ_(R)+ζ₀),  (4)

S _(ψ2)(t)=ε₂ cos(φ_(R)+2φ+ζ₂),  (5)

S _(ψ3)(t)=ε₃ cos(φ_(R)+3φ+ζ₃);  (6)

φ_(R) is the phase of a reference signal S_(R)(t) with dφ_(R)/dt=ω_(R)corresponding to 2π times the frequency difference of the measurementbeam and reference beam components of the input beam to theinterferometer; A₁ and ζ₁ are the amplitude and offset phase,respectively, of the primary component of the electrical interferencesignal; ε⁻¹, ε₀, ε₂, and ε₃ are the amplitudes for the cyclic errorterms; ζ⁻¹, ζ₀, ζ₂, and ζ₃ are the offset phases of the cyclic errorterms;

φ=4kL  (7)

for a plane mirror interferometer such as a HSPMI (which involves twopasses of the measurement beam to the measurement object); k is awavenumber corresponding to wavelength λ of beam 11; and L is thedifference between the one way physical length of the measurement pathand the one way physical length of the reference path of theinterferometer. Cyclic error amplitudes ε⁻¹, ε₀, ε₂, and ε₃ are muchless than the A₁, i.e. ≲( 1/50)A₁. An example of the frequencydifference ω_(R)/2π is 20 MHz.

Note that there is generally a set of cyclic error terms whose phasesare independent of φ_(R). This set of cyclic error terms has beenomitted from Equation (2) because they are eliminated by a high passfilter in converter/filter 52.

The factors relating to amplitudes ε_(p) and offset phases ζ_(p) of thecyclic error terms and the time dependent factors of the cyclic errorterms are generated using measured values of both S(t) and referencesignal S_(R)(t). The factors relating to amplitudes ε_(p) and offsetphases ζ_(p) are determined and the results transmitted to a table 40for subsequent use in generation of the cyclic error correction signalS_(ψ)(t). The time dependent factors of the cyclic error terms areobtained by application of simple discrete transforms based ontrigonometric identities and properties of conjugated quadratures ofsignals.

Optical reference signal 13 is detected by detector 14 to produce anelectrical reference signal. The optical reference signal can be derivedfrom a portion of the input beam to the interferometer. Alternatively,the electrical reference signal can be derived directly from the sourcethat introduces the heterodyne frequency splitting in the input beamcomponents (e.g., from the drive signal to an acousto-optical modulatorused to generate the heterodyne frequency splitting). The electricalreference signal is converted to a digital format and passed through ahigh pass filter in converter/filter 54 to produce reference signalS_(R)(t). Reference signal S_(R)(t) in digital format is written as

S _(R)(t)=A _(R) cos(φ_(R)+ζ_(R))  (8)

where A_(R) and ζ_(R) are the amplitude and offset phase, respectively,of the reference signal. The ADC conversion rate in 54 for S_(R)(t) isthe same as the ADC conversion rate in 52 for S(t). The quadraturesignal {tilde over (S)}_(R)(t) of S_(R)(t) written as

{tilde over (S)} _(R)(t)=A _(R) sin(φ_(R)+ζ_(R))  (9)

is generated by electronic processing using measured values of S_(R)(t)according to the formula

$\begin{matrix}{{{\overset{\sim}{S}}_{R}(t)} = {{\left( {\cot \; \omega_{R}\tau} \right){S_{R}\left( {t - {2\tau}} \right)}} - {\frac{\cos \; 2\omega_{R}\tau}{\sin \; \omega_{R}\tau}{S_{R}\left( {t - \tau} \right)}}}} & (10)\end{matrix}$

where 1/τ is the ADC conversion rate of reference signal S_(R)(t) in 54.For the example of a frequency difference for ω_(R)/2π=20 MHz and an ADCconversion rate 1/τ of 120 MHz, Equation (10) reduces to a particularlysimple form

$\begin{matrix}{{{\overset{\sim}{S}}_{R}(t)} = {{\frac{1}{\sqrt{3}}\left\lbrack {{S_{R}\left( {t - \tau} \right)} + {S_{R}\left( {t - {2\tau}} \right)}} \right\rbrack}.}} & (11)\end{matrix}$

Reference signal S_(R)(t) and quadrature signal {tilde over (S)}_(R)(t)are conjugated quadratures of reference signal S_(R)(t). The quadraturesignal {tilde over (S)}_(R)(t) is generated by processor 16 usingEquation (11) or Equation (10) as appropriate.

The quadrature signal S(t) of S(t) is generated by processor 56 usingthe same processing procedure as that described for the generation ofquadrature signal {tilde over (S)}_(R)(t). Accordingly, for the exampleof a ratio of 1/τ and ω_(R)/2π equal to 6,

$\begin{matrix}{\begin{matrix}{{\overset{\sim}{S}(t)} = {\frac{1}{\sqrt{3}}\left\lbrack {{S\left( {t - \tau} \right)} + {S\left( {t - {2\tau}} \right)}} \right\rbrack}} \\{= {{A_{0}\sin \; \left( {\phi_{R} + \phi + \zeta_{1}} \right)} + {{\overset{\sim}{S}}_{\psi}(t)}}}\end{matrix}{where}} & (12) \\\begin{matrix}{{{\overset{\sim}{S}}_{\psi}(t)} = {{ɛ_{- 1}{\sin \left( {\phi_{R} - \phi + \zeta_{- 1}} \right)}} + {ɛ_{0}{\sin \left( {\phi_{R} + \zeta_{0}} \right)}} +}} \\{{{ɛ_{2}{\sin \left( {\phi_{R} + {2\phi} + \zeta_{2}} \right)}} + {ɛ_{3}{{\sin \left( {\phi_{R} + {3\phi} + \zeta_{3}} \right)}.}}}}\end{matrix} & (13)\end{matrix}$

Equation (12) is valid when the stage slew rate is low, e.g., when φchanges insignificantly over the time period 2τ. Signal S(t) andquadrature signal {tilde over (S)}(t) are conjugated quadratures ofsignal S(t).

Notably, the integral relationship between 1/τ and ω_(R)/2π allows thegeneration of feed forward values S′(t) and {tilde over (S)}′(t) of S(t)and {tilde over (S)}(t), respectively, and S_(R)(t) and {tilde over(S)}′_(R)(t) of S_(R)(t) and {tilde over (S)}_(R)(t), respectively, areaccording to the formulae

S′(t)=S(t−6mτ),  (14)

S′(t)=S(t−6mτ),  (15)

S′ _(R)(t)=S _(R)(t−6mτ),  (16)

{tilde over (S)}′ _(R)(t)={tilde over (S)} _(R)(t−6mτ)  (17)

where m is an integer such that the error in the phases of feed forwardsignals with respect to corresponding phases of signals is less thanpredetermined values set by an end use application. In other words,prior values of the main interference signal and the reference signalcan be used to generate the quadrature signals and subsequent errorbasis functions. In other embodiments in which the ratio between 1/τ andω_(R)/2π is an integer different from 6, Equations (14)-(17) aremodified accordingly.

Using algebraic combinations of the signals S(t), {tilde over (S)}(t),S_(R)(t), and {tilde over (S)}_(R)(t), processing unit 10 generatescyclic error basis functions, which are sine and cosine functions thathave the same time-varying arguments as the cyclic error terms, and thenuses the cyclic error basis functions to project out respective cyclicerror coefficients from S(t) and {tilde over (S)}(t) by low-passfiltering (e.g., averaging). The cyclic error basis functions forS_(ψ0)(t), for example, are especially simple and correspond to thereference signal and its quadrature signal S_(R)(t), and {tilde over(S)}_(R)(t). In other words, to process the signals for informationabout the cyclic error term ε₀ cos(φ_(R)+ζ₀), signals S_(R)(t) and{tilde over (S)}_(R)(t) are used as time dependent factors in therepresentation of the cyclic error term ε₀ cos(φ_(R)+ζ₀).

To better understand the representation, it is beneficial to rewritecyclic error term ε₀ cos(φ_(R)+ζ₀) in terms of the time dependentfunctions cos(φ_(R)+ζ_(R)) and sin(φ_(R)+ζ_(R)) with the results

$\begin{matrix}{{{ɛ_{0}{\cos \left( {\phi_{R} + \zeta_{0}} \right)}} = {ɛ_{0}\begin{bmatrix}{{{\cos \left( {\zeta_{0} - \zeta_{R}} \right)}{\cos \left( {\phi_{R} + \zeta_{R}} \right)}} -} \\{{\sin \left( {\zeta_{0} - \zeta_{R}} \right)}{\sin \left( {\phi_{R} + \zeta_{R}} \right)}}\end{bmatrix}}},} & (18) \\{{ɛ_{0}{\sin \left( {\phi_{R} + \zeta_{0}} \right)}} = {{ɛ_{0}\begin{bmatrix}{{{\cos \left( {\zeta_{0} - \zeta_{R}} \right)}{\sin \left( {\phi_{R} + \zeta_{R}} \right)}} +} \\{{\sin \left( {\zeta_{0} - \zeta_{R}} \right)}{\cos \left( {\phi_{R} + \zeta_{R}} \right)}}\end{bmatrix}}.}} & (19)\end{matrix}$

Equations (18) and (19) can be rewritten as

ε₀ cos(φ_(R)+ζ₀)=[A ₀ cos(φ_(R)+ζ_(R))−B ₀ sin(φ_(R)+ζ_(R))],  (20)

ε₀ sin(φ_(R)+ζ₀)=[A ₀ sin(φ_(R)+ζ_(R))+B ₀ cos(φ_(R)+ζ_(R))]  (21)

where

A ₀=ε₀ cos(ζ₀−ζ_(R)),  (22)

B ₀=ε₀ sin(ζ₀−ζ_(R)).  (23)

Conjugated quadratures S(t) and {tilde over (S)}(t) and conjugatedquadratures S_(R)(t) and {tilde over (S)}_(R)(t) are transmitted toprocessor 20 wherein signals Σ₀(t) and {tilde over (Σ)}₀(t) aregenerated. Signals Σ₀(t) and {tilde over (Σ)}₀(t) are given by theequations

Σ₀(t)≡S(t)S _(R)(t)+{tilde over (S)}(t){tilde over (S)} _(R)(t),  (24)

{tilde over (Σ)}₀(t)≡−S(t){tilde over (S)} _(R)(t)+{tilde over (S)}(t)S_(R)(t).  (25)

Using properties of conjugated quadratures and certain trigonometricidentities, e.g., cos² γ+sin² γ=1, we have

$\begin{matrix}{{{\Sigma_{0}(t)} = {{A_{R}A_{0}} + {A_{R}\begin{bmatrix}{{A_{1}{\cos \left( {\phi + \zeta_{1} - \zeta_{R}} \right)}} +} \\{{ɛ_{- 1}{\cos \left( {{- \phi} + \zeta_{- 1} - \zeta_{R}} \right)}} +} \\{{ɛ_{2}{\cos \left( {{2\phi} + \zeta_{2} - \zeta_{R}} \right)}} +} \\{ɛ_{3}{\cos \left( {{3\phi} + \zeta_{3} - \zeta_{R}} \right)}}\end{bmatrix}}}},} & (26) \\{{{\overset{\sim}{\Sigma}}_{0}(t)} = {{A_{R}B_{0}} + {{A_{R}\begin{bmatrix}{{A_{1}{\sin \left( {\phi + \zeta_{1} - \zeta_{R}} \right)}} +} \\{{ɛ_{- 1}{\sin \left( {{- \phi} + \zeta_{- 1} - \zeta_{R}} \right)}} +} \\{{ɛ_{2}{\sin \left( {{2\phi} + \zeta_{2} - \zeta_{R}} \right)}} +} \\{ɛ_{3}{\sin \left( {{3\phi} + \zeta_{3} - \zeta_{R}} \right)}}\end{bmatrix}}.}}} & (27)\end{matrix}$

Notably, the generation of signals Σ₀(t) and {tilde over (Σ)}₀(t)project the coefficients associated with S_(ψ0)(t) to zero frequency,where low-pass filtering techniques can be used to determine them. Thus,signals Σ₀(t) and {tilde over (Σ)}₀(t) are transmitted to low passdigital filters in processor 24, e.g., low pass Butterworth filters,where coefficients A_(R)A₀ and A_(R)B₀ are determined.

For a Butterworth filter T_(n)(x) of order n, the corresponding outputsof the low pass digital filters for inputs Σ₀(t) and {tilde over(Σ)}₀(t) are

$\begin{matrix}{{{T_{n}\left\lbrack {\Sigma_{0}(t)} \right\rbrack} = {{A_{R}A_{0}} + {A_{R}\begin{bmatrix}{{A_{1}{O\left( \frac{\omega_{c}}{\omega_{D}} \right)}^{n}} + {ɛ_{- 1}{O\left( \frac{\omega_{c}}{\omega_{D}} \right)}^{n}} +} \\{{ɛ_{2}{O\left( \frac{\omega_{c}}{2\omega_{D}} \right)}^{n}} + {ɛ_{3}{O\left( \frac{\omega_{c}}{3\omega_{D}} \right)}^{n}}}\end{bmatrix}}}},} & (28) \\\begin{matrix}{{\Sigma_{2}(t)} \equiv {{S_{1} \cdot \Sigma_{1}} - {{\overset{\sim}{S}}_{1} \cdot {\overset{\sim}{\Sigma}}_{1}}}} \\{= {{A_{R}{A_{1}\begin{bmatrix}{{A_{1}{\cos \left( {\phi_{R} + {2\phi} + {2\zeta_{1}} - \zeta_{R}} \right)}} +} \\{{2ɛ_{- 1}{\cos \left( {\phi_{R} + \zeta_{1} + \zeta_{- 1} - \zeta_{R}} \right)}} +} \\{{2ɛ_{2}{\cos \left( {\phi_{R} + {3\phi} + \zeta_{1} + \zeta_{2} - \zeta_{R}} \right)}} +} \\{2ɛ_{3}{\cos \left( {\phi_{R} + {4\phi} + \zeta_{1} + \zeta_{3} - \zeta_{R}} \right)}}\end{bmatrix}}} +}} \\{{{A_{R}{O\left( {ɛ_{i}ɛ_{j}} \right)}},}}\end{matrix} & (29)\end{matrix}$

where O(x) denotes a term of the order of x, ω_(c) is the −3 dB angularcutoff frequency, and ω_(D)=dφ/dt.

The terms on the right hand sides of Equations (28) and (29) withfactors A_(R)A₁ are the sources of the largest errors and accordinglydetermine the specifications of n and the minimum ratio for ω_(D)/ω_(c)that can be used when the outputs of processor 24 are stored in table40. For a fourth order Butterworth filter, i.e., n=4, and a minimumratio for ω_(D)/ω_(c)=7, the error terms on the right hand side ofEquations (28) and (29) will generate errors that correspond to ≲0.010nm (3σ). When the stage is moving at a speed such that the correspondingDoppler shift frequency ω_(D) /2π is 10 to 100 times greater than thebandwidth of the stage servo control system and the requirement withrespect to ω_(D)/ω_(c) is satisfied, the outputs A_(R)A₀ and A_(R)B₀ ofthe low pass filters in processor 24 are stored in table 40 and inprocessors 26 and 28 under the control of signal 72 (from processor 70).

Notably, in this preferred embodiment, ω_(D) can vary by factors such as2 or more during the period associated with output values of A_(R)A₀ andA_(R)B₀ that are stored in table 40.

Quadratures S_(R) and {tilde over (S)}_(R) are transmitted to processor22 and the value for A_(R) ² is determined in processor 22 by aprocedure similar to that used in processor 20 and processor 24 usingthe formulae

T _(n) [S _(R)(t)·S _(R)(t)+{tilde over (S)}_(R)(t)·{tilde over(S)}_(R)(t)]=A _(R) ².  (30)

The value of the order n need only be for example 2. Values of A_(R) ²are transmitted to table 40 and stored under the control of signal 72.

Alternatively, detector 14, converter/filter 54, and processor 16 can bereplaced by a lookup table synchronized to the reference signal. Thiscan potentially reduce uncertainty caused by noise on the referencesignal. If the value of A_(R) ² is normalized to unity then someequations can be simplified and processor 22 can be removed.

The values for A_(R)A₀, A_(R)B₀, S(t), {tilde over (S)}_(R)(t) S_(R)(t),{tilde over (S)}_(R)(t), and A_(R) ² are transmitted to processor 26 andthe values of A_(R)A₀, A_(R)B₀, and A_(R) ² are stored in processor 26under the control of signal 72 for the generation of conjugatedquadratures S₁(t) and {tilde over (S)}₁(t) where

$\begin{matrix}\begin{matrix}{{S_{1}(t)} \equiv {{S(t)} - {\frac{\left( {A_{R}A_{0}} \right)}{A_{R}^{2}}S_{R}} + {\frac{\left( {A_{R}B_{0}} \right)}{A_{R}^{2}}{\overset{\sim}{S}}_{R}}}} \\{= {{A_{1}\cos \; \left( {\phi_{R} + \phi + \zeta_{1}} \right)} + {ɛ_{- 1}{\cos \left( {\phi_{R} - \phi + \zeta_{- 1}} \right)}} +}} \\{{{{ɛ_{2}{\cos \left( {\phi_{R} + {2\phi} + \zeta_{2}} \right)}} + {ɛ_{3}{\cos \left( {\phi_{R} + {3\phi} + \zeta_{3}} \right)}}},}}\end{matrix} & (31) \\\begin{matrix}{{{\overset{\sim}{S}}_{1}(t)} \equiv {{\overset{\sim}{S}(t)} - {\frac{\left( {A_{R}A_{0}} \right)}{A_{R}^{2}}{\overset{\sim}{S}}_{R}} - {\frac{\left( {A_{R}B_{0}} \right)}{A_{R}^{2}}S_{R}}}} \\{= {{A_{1}\sin \; \left( {\phi_{R} + \phi + \zeta_{1}} \right)} + {ɛ_{- 1}{\sin \left( {\phi_{R} - \phi + \zeta_{- 1}} \right)}} +}} \\{{{ɛ_{2}{\sin \left( {\phi_{R} + {2\phi} + \zeta_{2}} \right)}} + {ɛ_{3}{{\sin \left( {\phi_{R} + {3\phi} + \zeta_{3}} \right)}.}}}}\end{matrix} & (32)\end{matrix}$

The values for A_(R)A₀, A_(R)B₀, Σ₀(t), and {tilde over (Σ)}₀(t) aretransmitted to processor 28 and the values of A_(R)A₀ and A_(R)B₀ arestored in processor 28 under the control of signal 72 for the generationof conjugated quadratures Σ₁(t) and {tilde over (Σ)}₁(t) where

$\begin{matrix}\begin{matrix}{{\Sigma_{1}(t)} \equiv {{\Sigma_{0}(t)} - {A_{R}A_{0}}}} \\{{= {A_{R}\begin{bmatrix}{{A_{1}{\cos \left( {\phi + \zeta_{1} - \zeta_{R}} \right)}} + {ɛ_{- 1}{\cos \left( {{- \phi} + \zeta_{- 1} - \zeta_{R}} \right)}} +} \\{{ɛ_{2}{\cos \left( {{2\phi} + \zeta_{2} - \zeta_{R}} \right)}} + {ɛ_{3}{\cos \left( {{3\phi} + \zeta_{3} - \zeta_{R}} \right)}}}\end{bmatrix}}},}\end{matrix} & (33) \\\begin{matrix}{{{\overset{\sim}{\Sigma}}_{1}(t)} \equiv {{{\overset{\sim}{\Sigma}}_{0}(t)} - {A_{R}B_{0}}}} \\{= {{A_{R}\begin{bmatrix}{{A_{1}{\sin \left( {\phi + \zeta_{1} - \zeta_{R}} \right)}} + {ɛ_{- 1}{\sin \left( {{- \phi} + \zeta_{- 1} - \zeta_{R}} \right)}} +} \\{{ɛ_{2}{\sin \left( {{2\phi} + \zeta_{2} - \zeta_{R}} \right)}} + {ɛ_{3}{\sin \left( {{3\phi} + \zeta_{3} - \zeta_{R}} \right)}}}\end{bmatrix}}.}}\end{matrix} & (34)\end{matrix}$

Signals S_(R)(t), {tilde over (S)}_(R)(t), Σ₁(t), and {tilde over(Σ)}₁(t) are transmitted to processor 30 for the generation ofconjugated quadratures Σ⁻¹(t) and {tilde over (Σ)}⁻¹(t) where

$\begin{matrix}\begin{matrix}{{\Sigma_{- 1}(t)} \equiv {{\Sigma_{1}S_{R}} + {{\overset{\sim}{\Sigma}}_{1}{\overset{\sim}{S}}_{R}}}} \\{{= {A_{R}^{2}\begin{bmatrix}{{ɛ_{- 1}{\cos \left( {\phi_{R} + \phi - \zeta_{- 1} + {2\zeta_{R}}} \right)}} +} \\{{A_{1}{\cos \left( {\phi_{R} - \phi - \zeta_{1} + {2\zeta_{R}}} \right)}} +} \\{{ɛ_{2}{\cos \left( {\phi_{R} - {2\phi} - \zeta_{2} + {2\zeta_{R}}} \right)}} +} \\{ɛ_{3}{\cos \left( {\phi_{R} - {3\phi} - \zeta_{3} + {2\zeta_{R}}} \right)}}\end{bmatrix}}},}\end{matrix} & (35) \\\begin{matrix}{{{\overset{\sim}{\Sigma}}_{- 1}(t)} \equiv {{\Sigma_{1}{\overset{\sim}{S}}_{R}} - {{\overset{\sim}{\Sigma}}_{1}S_{R}}}} \\{= {{A_{R}^{2}\begin{bmatrix}{{ɛ_{- 1}{\sin \left( {\phi_{R} + \phi - \zeta_{- 1} + {2\zeta_{R}}} \right)}} +} \\{{A_{1}{\sin \left( {\phi_{R} - \phi - \zeta_{1} + {2\zeta_{R}}} \right)}} +} \\{{ɛ_{2}{\sin \left( {\phi_{R} - {2\phi} - \zeta_{2} + {2\zeta_{R}}} \right)}} +} \\{ɛ_{3}{\sin \left( {\phi_{R} - {3\phi} - \zeta_{3} + {2\zeta_{R}}} \right)}}\end{bmatrix}}.}}\end{matrix} & (36)\end{matrix}$

Signals Σ⁻¹(t) and {tilde over (Σ)}⁻¹(t) correspond to the cyclic errorbasis functions for S_(ψ−1)(t) in that the leading terms of Σ⁻¹(t) and{tilde over (Σ)}⁻¹(t) are sinusoids with the same time-varying argumentas that of S_(ψ−1)(t).

Coefficients A_(R) ² A₁A⁻¹ and −A_(R) ² A₁B⁻¹ are next determinedthrough digital low pass filters, e.g., low pass Butterworth filters, inprocessor 32 where

A ⁻¹≡ε⁻¹ cos(ζ⁻¹+ζ₁−2ζ_(R)),  (37)

B ⁻¹≡ε⁻¹ sin(ζ⁻¹+ζ₁−2ζ_(R)).  (38)

The input signals for the digital filters are Σ₄(t) and {tilde over(Σ)}₄(t). Input signals Σ₄(t) and {tilde over (Σ)}₄(t) are generated inprocessor 32 using signals S₁, {tilde over (S)}₁, Σ⁻¹(t), and {tildeover (Σ)}⁻¹(t) according to the formulae

Σ₄(t)=[S ₁(t)Σ⁻¹(t)+{tilde over (S)} ₁(t){tilde over (Σ)}⁻¹(t)],  (39)

{tilde over (Σ)}₄(t)=[S ₁(t){tilde over (Σ)}⁻¹(t)−{tilde over (S)}₁(t)Σ⁻¹(t)].  (40)

Equations (39) and (40) are written in terms of A⁻¹ and B⁻¹ usingEquations (37) and (38) as

$\begin{matrix}\begin{matrix}{\Sigma_{4} = {{2A_{R}^{2}A_{1}A_{- 1}} +}} \\{{{A_{R}^{2}{A_{1}\begin{bmatrix}{{A_{1}{\cos \left( {{{- 2}\phi} - \zeta_{1} - \zeta_{1} + {2\zeta_{R}}} \right)}} +} \\{{2ɛ_{2}{\cos \left( {{{- 3}\phi} - \zeta_{1} - \zeta_{2} + {2\zeta_{R}}} \right)}} +} \\{2ɛ_{3}{\cos \left( {{{- 4}\phi} - \zeta_{1} - \zeta_{3} + {2\zeta_{R}}} \right)}}\end{bmatrix}}} +}} \\{{{A_{R}^{2}{O\left( {ɛ_{i}ɛ_{j}} \right)}},}}\end{matrix} & (41) \\\begin{matrix}{{\overset{\sim}{\Sigma}}_{4} = {{{- 2}A_{R}^{2}A_{1}B_{- 1}} +}} \\{{{A_{R}^{2}{A_{1}\begin{bmatrix}{{A_{1}{\sin \left( {{{- 2}\phi} - \zeta_{1} - \zeta_{1} + {2\zeta_{R}}} \right)}} +} \\{{2ɛ_{2}{\sin \left( {{{- 3}\phi} - \zeta_{1} - \zeta_{2} + {2\zeta_{R}}} \right)}} +} \\{2ɛ_{3}{\sin \left( {{{- 4}\phi} - \zeta_{1} - \zeta_{3} + {2\zeta_{R}}} \right)}}\end{bmatrix}}} +}} \\{{A_{R}^{2}{{O\left( {ɛ_{i}ɛ_{j}} \right)}.}}}\end{matrix} & (42)\end{matrix}$

Signals Σ₄(t) and {tilde over (Σ)}₄(t) are sent to low pass digitalfilters in processor 32, e.g., low pass Butterworth filters, wherecoefficients 2A_(R) ²A₁A⁻¹ and −2A_(R) ²A₁B⁻¹ are determined. For aButterworth filter T_(n)(x) of order n, the corresponding outputs of thelow pass digital filters for inputs Σ₄(t) and {tilde over (Σ)}₄(t) are

$\begin{matrix}{{{T_{n}\left\lbrack {\Sigma_{4}(t)} \right\rbrack} = {{2A_{R}^{2}A_{1}A_{- 1}} + {A_{R}^{2}{A_{1}\left\lbrack {{A_{1}{O\left( \frac{\omega_{c}}{2\omega_{D}} \right)}^{n}} + {2ɛ_{2}{O\left( \frac{\omega_{c}}{3\omega_{D}} \right)}^{n}} + {2ɛ_{3}{O\left( \frac{\omega_{c}}{4\omega_{D}} \right)}^{n}}} \right\rbrack}}}},} & (43) \\{{T_{n}\left\lbrack {{\overset{\sim}{\Sigma}}_{4}(t)} \right\rbrack} = {{{- 2}A_{R}^{2}A_{1}B_{- 1}} + {A_{R}^{2}{{A_{1}\left\lbrack {{A_{1}{O\left( \frac{\omega_{c}}{2\omega_{D}} \right)}^{n}} + {2ɛ_{2}{O\left( \frac{\omega_{c}}{3\omega_{D}} \right)}^{n}} + {2ɛ_{3}{O\left( \frac{\omega_{c}}{4\omega_{D}} \right)}^{n}}} \right\rbrack}.}}}} & (44)\end{matrix}$

The terms on the right hand sides of Equations (43) and (44) withfactors A_(R) ² A₁ ² are the sources of the largest errors andaccordingly determine the specifications of n and the minimum ratio forω_(D)/ω_(c) that can be used when the outputs of processor 32 are storedin table 40. For a fourth order Butterworth filter, i.e., n=4, and aminimum ratio for ω_(D)/ω_(c)=3.5, the error terms on the right handside of Equations (43) and (44) will generate errors that correspond to≲0.010 nm (3σ). The outputs 2A_(R) ² A₁A⁻¹ and −2A_(R) ²A₁B⁻¹ of lowpass filters of processor 32 are divided by 2 to generate A_(R) ²A₁A⁻¹and −A_(R) ²A₁B⁻¹ as the outputs of processor 32. When the stage ismoving at a speed such that the corresponding Doppler shift frequencyω_(D)/2π is 10 to 100 times greater than the bandwidth of the stageservo control system and the requirement with respect to ω_(D)/ω_(c) issatisfied, the outputs A_(R) ²A₁A⁻¹ and −A_(R) ²A₁B⁻¹ of processor 32are stored in table 40 and in processor 34 under the control of signal72.

Signals S_(1(t) and {tilde over (S)}) ₁(t) are transmitted to processor30 for the generation of conjugated quadratures Σ₂(t) and {tilde over(Σ)}₂(t) where

$\begin{matrix}\begin{matrix}{{\Sigma_{2}(t)} \equiv {{S_{1} \cdot \Sigma_{1}} - {{\overset{\sim}{S}}_{1} \cdot {\overset{\sim}{\Sigma}}_{1}}}} \\{= {{A_{R}{A_{1}\begin{bmatrix}{{A_{1}{\cos \left( {\phi_{R} + {2\phi} + {2\zeta_{1}} - \zeta_{R}} \right)}} +} \\{{2ɛ_{- 1}{\cos \left( {\phi_{R} + \zeta_{1} + \zeta_{- 1} - \zeta_{R}} \right)}} +} \\{{2ɛ_{2}{\cos \left( {\phi_{R} + {3\phi} + \zeta_{1} + \zeta_{2} - \zeta_{R}} \right)}} +} \\{2ɛ_{3}{\cos \left( {\phi_{R} + {4\phi} + \zeta_{1} + \zeta_{3} - \zeta_{R}} \right)}}\end{bmatrix}}} +}} \\{{{A_{R}{O\left( {ɛ_{i}ɛ_{j}} \right)}},}}\end{matrix} & (45) \\\begin{matrix}{{{\overset{\sim}{\Sigma}}_{2}(t)} \equiv {{S_{1} \cdot {\overset{\sim}{\Sigma}}_{1}} + {{\overset{\sim}{S}}_{1} \cdot \Sigma_{1}}}} \\{= {{A_{R}{A_{1}\begin{bmatrix}{{A_{1}{\sin \left( {\phi_{R} + {2\phi} + {2\zeta_{1}} - \zeta_{R}} \right)}} +} \\{{2ɛ_{- 1}{\sin \left( {\phi_{R} + \zeta_{1} + \zeta_{- 1} - \zeta_{R}} \right)}} +} \\{{2ɛ_{2}{\sin \left( {\phi_{R} + {3\phi} + \zeta_{1} + \zeta_{2} - \zeta_{R}} \right)}} +} \\{2ɛ_{3}{\sin \left( {\phi_{R} + {4\phi} + \zeta_{1} + \zeta_{3} - \zeta_{R}} \right)}}\end{bmatrix}}} +}} \\{{A_{R}{{O\left( {ɛ_{i}ɛ_{j}} \right)}.}}}\end{matrix} & (46)\end{matrix}$

Signals Σ₂(t) and {tilde over (Σ)}₂(t) correspond to the cyclic errorbasis functions for S_(ψ2)(t) in that the leading terms of Σ₂(t) and{tilde over (Σ)}₂(t) are sinusoids with the same time-varying argumentas that of S_(ψ2)(t).

Signals S₁, {tilde over (S)}₁, Σ₁(t), {tilde over (Σ)}₁(t), Σ₂(t), and{tilde over (Σ)}₂(t) and coefficients A_(R) ²A₁A⁻¹, and are −A_(R)²A₁B⁻¹ transmitted to processor 34 and coefficients A_(R) ²A₁A⁻¹, and−A_(R) ²A₁B⁻¹ stored in processor 34 under the control of signal 72 forgeneration of conjugated quadratures Σ₃(t) and {tilde over (Σ)}₃(t)where

$\begin{matrix}\begin{matrix}{\mspace{20mu} {{\Sigma_{3}(t)} \equiv {{\Sigma_{1} \cdot \Sigma_{2}} - {{\overset{\sim}{\Sigma}}_{1} \cdot {\overset{\sim}{\Sigma}}_{2}} + {3\left\lbrack {{A_{R}^{2}A_{1}A_{- 1}S_{1}} - {A_{R}^{2}A_{1}B_{- 1}{\overset{\sim}{S}}_{1}}} \right\rbrack}}}} \\{= {{A_{R}^{2}{A_{1}^{2}\begin{bmatrix}{{3ɛ_{- 1}{\cos \left( {\phi_{R} + \phi + {2\zeta_{1}} + \zeta_{- 1} - {2\zeta_{R}}} \right)}} +} \\{{A_{1}{\cos \left( {\phi_{R} + {3\phi} + {2\zeta_{1}} + \zeta_{1} - {2\zeta_{R}}} \right)}} +} \\{{3ɛ_{2}{\cos \left( {\phi_{R} + {4\phi} + {2\zeta_{1}} + \zeta_{2} - {2\zeta_{R}}} \right)}} +} \\{3ɛ_{3}{\cos \left( {\phi_{R} + {5\phi} + {2\zeta_{1}} + \zeta_{3} - {2\zeta_{R}}} \right)}}\end{bmatrix}}} -}} \\{{{{3\left\lbrack {{A_{R}^{2}A_{1}A_{- 1}S_{1}} - {A_{R}^{2}A_{1}B_{- 1}{\overset{\sim}{S}}_{1}}} \right\rbrack} + {A_{R}^{2}A_{1}{O\left( {ɛ_{i}ɛ_{j}} \right)}} + \ldots}\mspace{11mu},}}\end{matrix} & (47) \\\begin{matrix}{\mspace{20mu} {{\Sigma_{3}(t)} = {{A_{R}^{2}{A_{1}^{2}\begin{bmatrix}{{A_{1}{\cos \left( {\phi_{R} + {3\phi} + {3\zeta_{1}} + {2\zeta_{R}}} \right)}} +} \\{{3ɛ_{2}{\cos \left( {\phi_{R} + {4\phi} + {2\zeta_{1}} + \zeta_{2} - {2\zeta_{R}}} \right)}} +} \\{3ɛ_{3}{\cos \left( {\phi_{R} + {5\phi} + {2\zeta_{1}} + \zeta_{3} - {2\zeta_{R}}} \right)}}\end{bmatrix}}} +}}} \\{{{{A_{R}^{2}A_{1}{O\left( {ɛ_{i}ɛ_{j}} \right)}} + \ldots}\mspace{11mu},}}\end{matrix} & (48) \\\begin{matrix}{\mspace{20mu} {{{\overset{\sim}{\Sigma}}_{3}(t)} \equiv {{\Sigma_{1} \cdot {\overset{\sim}{\Sigma}}_{2}} + {{\overset{\sim}{\Sigma}}_{1} \cdot \Sigma_{2}} - {3\left\lbrack {{A_{R}^{2}A_{1}A_{- 1}{\overset{\sim}{S}}_{1}} - {A_{R}^{2}A_{1}B_{- 1}S_{1}}} \right\rbrack}}}} \\{= {{A_{R}^{2}{A_{1}^{2}\begin{bmatrix}{{3ɛ_{- 1}{\sin \left( {\phi_{R} + \phi + {2\zeta_{1}} + \zeta_{- 1} - {2\zeta_{R}}} \right)}} +} \\{{A_{1}{\sin \left( {\phi_{R} + {3\phi} + {2\zeta_{1}} + \zeta_{1} - {2\zeta_{R}}} \right)}} +} \\{{3ɛ_{2}{\sin \left( {\phi_{R} + {4\phi} + {2\zeta_{1}} + \zeta_{2} - {2\zeta_{R}}} \right)}} +} \\{3ɛ_{3}{\sin \left( {\phi_{R} + {5\phi} + {2\zeta_{1}} + \zeta_{3} - {2\zeta_{R}}} \right)}}\end{bmatrix}}} -}} \\{{{3\left\lbrack {{A_{R}^{2}A_{1}A_{- 1}{\overset{\sim}{S}}_{1}} + {A_{R}^{2}A_{1}B_{- 1}S_{1}}} \right\rbrack} + {A_{R}^{2}A_{1}{O\left( {ɛ_{i}ɛ_{j}} \right)}} + \ldots}}\end{matrix} & (49) \\{{{\overset{\sim}{\Sigma}}_{3}(t)} = {{A_{R}^{2}{A_{1}^{2}\begin{bmatrix}{{A_{1}{\sin \left( {\phi_{R} + {3\phi} + {3\zeta_{1}} - {2\zeta_{R}}} \right)}} +} \\{{3ɛ_{2}{\sin \left( {\phi_{R} + {4\phi} + {2\zeta_{1}} + \zeta_{2} - {2\zeta_{R}}} \right)}} +} \\{3ɛ_{3}{\sin \left( {\phi_{R} + {5\phi} + {2\zeta_{1}} + \zeta_{3} - {2\zeta_{R}}} \right)}}\end{bmatrix}}} + {A_{R}^{2}A_{1}{O\left( {ɛ_{i}ɛ_{j}} \right)}} + {\ldots \mspace{11mu}.}}} & (50)\end{matrix}$

Signals Σ₃(t) and {tilde over (Σ)}₃(t) correspond to the cyclic errorbasis functions for S_(ψ3)(t) in that the leading terms of Σ₃(t) and{tilde over (Σ)}₃(t) are sinusoids with the same time-varying argumentas that of S_(ψ3)(t).

Coefficients A_(R)A₁ ²A₂ and −A_(R)A₁ ²B₂ (where A₂ and B₂ are cyclicerror coefficients for S_(ψ2) and are given explicitly by Equations (67)and (68), respectively, further below) are next determined throughdigital low pass filters, e.g., low pass Butterworth filters, inprocessor 38. The input signals for the digital filters are Σ₅(t) and{tilde over (Σ)}₅(t), respectively. The input signals are generated inprocessor 38 using signals S₁, {tilde over (S)}₁, Σ₂(t), and {tilde over(Σ)}₂(t) according to the formulae

Σ₅(t)≡[S ₁(t)Σ₂(t)+{tilde over (S)} ₁(t){tilde over (Σ)}₂(t)],  (51)

{tilde over (Σ)}₅(t)≡[S ₁(t){tilde over (Σ)}₂(t)−{tilde over (S)}₁(t)Σ₂(t)].  (52)

The expansions of Σ₅(t) and {tilde over (Σ)}₅(t), given by Equations(51) and (52), respectively, in terms of cyclic error and non-cyclicerror terms are

$\begin{matrix}{{{\Sigma_{5}(t)} = {{A_{R}A_{1}^{2}A_{2}} + {A_{R} A_{1}^{2} \begin{Bmatrix}\begin{matrix}{{ɛ_{- 1}\left\lbrack {{2{\cos \left( {{- \phi} + \zeta_{- 1} - \zeta_{R}} \right)}} + {\cos \left( {{3\phi} + {2\zeta_{1}} - \zeta_{- 1} - \zeta_{R}} \right)}} \right\rbrack} +} \\{{A_{1}{\cos \left( {\phi + \zeta_{1} - \zeta_{R}} \right)}} + {2ɛ_{2}{\cos \left( {{2\phi} + \zeta_{2} - \zeta_{R}} \right)}} +}\end{matrix} \\{ɛ_{3}\left\lbrack {{2{\cos \left( {{3\phi} + \zeta_{3} - \zeta_{R}} \right)}} + {\cos \left( {{- \phi} + {2\zeta_{1}} - \zeta_{3} - \zeta_{R}} \right)}} \right\rbrack}\end{Bmatrix}} + {A_{R}A_{1}{O\left( ɛ_{i}^{2} \right)}} + \ldots}}\mspace{11mu},} & (53) \\{{{{\overset{\sim}{\Sigma}}_{5}(t)} = {{{- A_{R}}A_{1}^{2}B_{2}} + {A_{R}A_{1}^{2}\begin{Bmatrix}{{ɛ_{- 1}\left\lbrack {{2{\sin \left( {{- \phi} + \zeta_{- 1} - \zeta_{R}} \right)}} + {\sin \left( {{3\phi} + {2\zeta_{1}} - \zeta_{- 1} - \zeta_{R}} \right)}} \right\rbrack} +} \\{{A_{1}{\sin \left( {\phi + \zeta_{1} - \zeta_{R}} \right)}} + {2ɛ_{2}{\sin \left( {{2\phi} + \zeta_{2} - \zeta_{R}} \right)}} +} \\{ɛ_{3}\left\lbrack {{2{\sin \left( {{3\phi} + \zeta_{3} - \zeta_{R}} \right)}} + {\sin \left( {{- \phi} + {2\zeta_{1}} - \zeta_{3} - \zeta_{R}} \right)}} \right\rbrack}\end{Bmatrix}} + {A_{R}A_{1}{O\left( ɛ_{i}^{2} \right)}} + \ldots}}\mspace{11mu},} & (54)\end{matrix}$

where A₂ and B₂ are given by Equations (67) and (68), respectively,shown further below.

Signals Σ₅(t) and {tilde over (Σ)}₅(t) are sent to low pass digitalfilters in processor 38, e.g., low pass Butterworth filters, wherecoefficients A_(R)A₁ ²A₂ and −A_(R)A₁ ²B₂ are determined. For aButterworth filter T_(n)(x) of order n, the corresponding outputs of thelow pass digital filters for inputs Σ₅(t) and {tilde over (Σ)}₅(t) are

$\begin{matrix}{{{T_{n}\left\lbrack {\Sigma_{5}(t)} \right\rbrack} = {{A_{R}A_{1}^{2}A_{2}} + {A_{R}{A_{1}^{2}\begin{bmatrix}{{A_{1}{O\left( \frac{\omega_{c}}{\omega_{D}} \right)}^{n}} + {2ɛ_{- 1}{O\left( \frac{\omega_{c}}{\omega_{D}} \right)}^{n}} + {ɛ_{- 1}{O\left( \frac{\omega_{c}}{3\omega_{D}} \right)}^{n}} +} \\{{2ɛ_{2}{O\left( \frac{\omega_{c}}{2\omega_{D}} \right)}^{n}} + {2ɛ_{3}{O\left( \frac{\omega_{c}}{3\omega_{D}} \right)}^{n}} + {ɛ_{2}{O\left( \frac{\omega_{c}}{\omega_{D}} \right)}^{n}}}\end{bmatrix}}}}},} & (55) \\{\mspace{20mu} {A_{R}^{2}{A_{1}^{4}.}}} & (56)\end{matrix}$

The terms on the right hand sides of Equations (55) and (56) withfactors A_(R)A₁ ³ are the sources of the largest errors and accordinglydetermine the specifications of n and the minimum ratio for ω_(D)/ω_(c)that can be used when the outputs of processor 38 are stored in table40. For a fourth order Butterworth filter, i.e., n=4, and a minimumratio for ω_(D)/ω_(c)=7, the error terms on the right hand side ofEquations (55) and (56) will generate errors that correspond to ≲0.010nm (3σ). The outputs A_(R)A₁ ²A₂ and −A_(R)A₁ ²B₂ of low pass filters ofprocessor 38 are the outputs of processor 38. When the stage is movingat a speed such that the corresponding Doppler shift frequency ω_(D) /2πis 10 to 100 times greater than the bandwidth of the stage servo controlsystem and the requirement with respect to ω_(D)/ω_(c) is satisfied, theoutputs A_(R)A₁ ²A₂ and −A_(R)A₁ ²B₂ of processor 38 are stored in table40 under the control of signal 72.

Coefficients A_(R) ²A₁ ³A₃ and −A_(R) ²A₁ ³B₃ (where A₃ and B₃ arecyclic error coefficients for S_(ψ3) and are given explicitly byEquations (69) and (70), respectively, further below) are nextdetermined through a digital low pass filter, e.g., a low passButterworth filter, in processor 36. The input signals for the digitalfilters are Σ₆(t) and {tilde over (Σ)}₆(t), respectively. The inputsignals are generated in processor 36 using signals S₁, {tilde over(S)}₁, Σ₃(t), and {tilde over (Σ)}₃(t) according to the formulae

Σ₆(t)=[S ₁(t)Σ₃(t)+{tilde over (S)} ₁(t){tilde over (Σ)}₃(t)],  (57)

{tilde over (Σ)}₆(t)=[S ₁(t){tilde over (Σ)}₃(t)−{tilde over (S)}₁(t)Σ₃(t)].  (58)

The expansions of Σ₆(t) and {tilde over (Σ)}₆(t) given by Equations (57)and (58), respectively, in terms of cyclic error and non-cyclic errorterms are

$\begin{matrix}{{{\Sigma_{6}(t)} = {{A_{R}^{2}A_{1}^{3}A_{3}} + {A_{R}^{2} {A_{1}^{3}\left\lbrack \begin{matrix}{{ɛ_{- 1}{\cos \left( {{4\phi} - \zeta_{- 1} + {3\zeta_{1}} - {2\zeta_{R}}} \right)}} +} \\{{A_{1}{\cos \left( {{2\phi} - \zeta_{1} + {3\zeta_{1}} - {2\zeta_{R}}} \right)}} +} \\{{ɛ_{2}{\cos \left( {\phi - \zeta_{2} + {3\zeta_{1}} - {2\zeta_{R}}} \right)}} + {3ɛ_{2}{\cos \left( {{3\phi} + \zeta_{1} + \zeta_{2} - {2\zeta_{R}}} \right)}} +} \\{3ɛ_{3}{\cos \left( {{4\phi} + \zeta_{1} + \zeta_{3} - {2\zeta_{R}}} \right)}}\end{matrix} \right\rbrack}} + {A_{R}^{2}A_{1}^{2}{O\left( ɛ_{i}^{2} \right)}} + \ldots}}\mspace{11mu},} & (59) \\{\mspace{20mu} {ɛ_{- 1}.}} & (60)\end{matrix}$

where A₃ and B₃ are given by Equations (69) and (70), respectively.

Signals Σ₆(t) and {tilde over (Σ)}₆(t) are sent to low pass digitalfilters in processor 36, e.g., low pass Butterworth filters, wherecoefficients A_(R) ² A₁ ³ A₃ and −A_(R) ² A₁ ³B₃ are determined. For aButterworth filter T_(n)(x) of order n, the corresponding outputs of thelow pass digital filters for inputs Σ₆(t) and {tilde over (Σ)}₆(t) are

$\begin{matrix}{{{T_{n}\left\lbrack {\Sigma_{6}(t)} \right\rbrack} = {{A_{R}^{2}A_{1}^{3}A_{3}} + {A_{R}^{2}{A_{1}^{3}\begin{bmatrix}{{A_{1}{O\left( \frac{\omega_{c}}{2\omega_{D}} \right)}^{n}} + {ɛ_{- 1}{O\left( \frac{\omega_{c}}{4\omega_{D}} \right)}^{n}} +} \\{{ɛ_{2}{O\left( \frac{\omega_{c}}{\omega_{D}} \right)}^{n}} + {3ɛ_{2}{O\left( \frac{\omega_{c}}{3\omega_{D}} \right)}^{n}} + {3ɛ_{3}{O\left( \frac{\omega_{c}}{4\omega_{D}} \right)}^{n}}}\end{bmatrix}}}}},} & (61) \\{{T_{n}\left\lbrack {\Sigma_{6}(t)} \right\rbrack} = {{{- A_{R}^{2}}A_{1}^{3}B_{3}} + {A_{R}^{2}{{A_{1}^{3}\begin{bmatrix}{{A_{1}{O\left( \frac{\omega_{c}}{2\omega_{D}} \right)}^{n}} + {ɛ_{- 1}{O\left( \frac{\omega_{c}}{4\omega_{D}} \right)}^{n}} +} \\{{ɛ_{2}{O\left( \frac{\omega_{c}}{\omega_{D}} \right)}^{n}} + {3ɛ_{2}{O\left( \frac{\omega_{c}}{3\omega_{D}} \right)}^{n}} + {3ɛ_{3}{O\left( \frac{\omega_{c}}{4\omega_{D}} \right)}^{n}}}\end{bmatrix}}.}}}} & (62)\end{matrix}$

The terms on the right hand sides of Equations (61) and (62) withfactors A_(R) ² A₁ ⁴ are the sources of the largest errors andaccordingly determined the specifications of n and the minimum ratio forω_(D)/ω_(c) that can be used when the outputs of processor 36 are storedin table 40. For a fourth order Butterworth filter, i.e., n=4, and aminimum ratio for ω_(D)/(c=3.5, the error terms on the right hand sideof Equations (61) and (62) will generate errors that correspond to≲0.010 nm (3σ). The outputs A_(R) ² A₁ ³A₃ and −A_(R) ²A₁ ³B₃ of lowpass filters of processor 36 are the outputs of processor 36. When thestage is moving at a speed such that the corresponding Doppler shiftfrequency ω_(D) /2π is 10 to 100 times greater than the bandwidth of thestage servo control system and the requirement with respect toω_(D)/ω_(c) is satisfied, the outputs A_(R) ²A₁ ³A₃ and −A_(R) ²A₁ ³B₃of processor 36 are stored in table 40 under the control of signal 72.

Finally, quadratures S and S are transmitted from processors 52 and 56respectively to processor 18 for the purpose of determining a value forA₁ ². First, a signal S(t)S(t)+{tilde over (S)}(t){tilde over (S)}(t) isgenerated where

$\begin{matrix}{{{{S(t)}{S(t)}} + {{\overset{\sim}{S}(t)}{\overset{\sim}{S}(t)}}} = {A_{1}^{2} + \left\lbrack {ɛ_{- 1}^{2} + ɛ_{0}^{2} + ɛ_{2}^{2} + ɛ_{3}^{2}} \right\rbrack + {2A_{1}ɛ_{- 1}{\cos \left( {{2\phi} + \zeta_{1} - \zeta_{- 1}} \right)}} + {2A_{1}ɛ_{0}{\cos \left( {\phi + \zeta_{1} - \zeta_{0}} \right)}} + {2A_{1}ɛ_{2}{\cos \left( {{- \phi} + \zeta_{1} - \zeta_{2}} \right)}} + {2A_{1}ɛ_{3}{\cos \left( {{{- 2}\phi} + \zeta_{1} - \zeta_{3}} \right)}} + {{O\left( {ɛ_{i}ɛ_{j}} \right)}.}}} & (63)\end{matrix}$

The signal of Equation (63) is sent to a low pass digital filter inprocessor 18, e.g., a low pass Butterworth filter, where the coefficientA₁ ² is determined. For a Butterworth filter T_(n)(x) of order n, thecorresponding outputs of the low pass digital filter for the signal ofEquation (63) is

$\begin{matrix}{{T_{n}\left\lbrack {{{S(t)} \cdot {S(t)}} + {{\overset{\sim}{S}(t)} \cdot {\overset{\sim}{S}(t)}}} \right\rbrack} = {A_{1}^{2} + \left\lbrack {ɛ_{- 1}^{2} + ɛ_{0}^{2} + ɛ_{2}^{2} + ɛ_{3}^{2}} \right\rbrack + {{A_{1}\begin{bmatrix}{{2ɛ_{- 1}{O\left( \frac{\omega_{c}}{2\omega_{D}} \right)}^{n}} + {2ɛ_{0}{O\left( \frac{\omega_{c}}{\omega_{D}} \right)}^{n}} +} \\{{2ɛ_{2}{O\left( \frac{\omega_{c}}{\omega_{D}} \right)}^{n}} + {2ɛ_{3}{O\left( \frac{\omega_{c}}{2\omega_{D\;}} \right)}^{n}}}\end{bmatrix}}.}}} & (64)\end{matrix}$

The accuracy required for the determination of A₁ ² is approximately0.5% in order to limit errors generated in the computation of cyclicerror signals S_(ψj) to ≲0.010 nm (3σ). Therefore the error terms ε⁻¹ ²,ε₀ ², ε₂ ², and ε₂ ² on the right hand side of Equation (64) arenegligible. The terms on the right hand side of Equation (64) of theform

${O\left( \frac{\omega_{c}}{\omega_{D}} \right)}^{n}$

are the sources of the largest Doppler shift frequency dependent errorsand accordingly determine the specifications of n and the minimum ratiofor ω_(D)/ω_(c) that can be used when the output of processor 18 isstored in table 40. For a second order Butterworth filter, i.e., n=2,and a minimum ratio for ω_(D)/ω_(c)=3.5, the Doppler shift frequencydependent error terms on the right hand side of Equation (64) willgenerate errors that correspond to ≲0.010 nm (3σ). The output A₁ ² ofthe low pass filter of processor 18 is the output of processor 18. Whenthe stage is moving at a speed such that the corresponding Doppler shiftfrequency ω_(D)/2π is 10 to 100 times greater than the bandwidth of thestage servo control system and the requirement with respect toω_(D)/ω_(c) is satisfied, the output A₁ ² of processor 18 is stored intable 40 under the control of signal 72.

Referring now to FIG. 4 b, processor 60 generates the compensating errorsignal S_(ψ). With respect to generating signal S_(ψ), it is beneficialto rewrite the ε⁻¹, ε₀, ε₂, and ε₃ cyclic error terms of S_(ψ) in termsof the highest order time dependent terms of Σ⁻¹(t), {tilde over(Σ)}⁻¹(t), S_(R), {tilde over (S)}_(R), Σ₂(t), {tilde over (Σ)}₂(t),Σ₃(t), and {tilde over (Σ)}₃(t), i.e., cos(φ_(R)−φ−ζ₁+2ζ_(R)),sin(φ_(R)−φ−ζ₁+2ζ_(R)), cos(φ_(R)+ζ_(R)), sin(φ_(R)+ζ_(R)),cos(φ_(R)+2φ+2ζ₁−ζ_(R)), sin(φ_(R)+2φ−2ζ₁+2ζ₁−ζ_(R)),cos(φ_(R)+3φ+3ζ₁−2ζ_(R)), and sin(φ_(R)+3φ+3ζ₁−2ζ_(R)) as

$\begin{matrix}{{S_{\psi}( t)} = {\left\lbrack \begin{matrix}{ɛ_{- 1}{\cos \left( {\zeta_{1} + \zeta_{- 1} - {2\zeta_{R}}} \right)}{\cos \left( {\phi_{R} - \phi - \zeta_{1} + {2\zeta_{R}}} \right)}} \\{{- ɛ_{- 1}}{\sin \left( {\zeta_{1} + \zeta_{- 1} - {2\zeta_{R}}} \right)}{\sin \left( {\phi_{R} - \phi - \zeta_{1} + {2\zeta_{R}}} \right)}}\end{matrix} \right\rbrack + \mspace{59mu} \left\lbrack \begin{matrix}{ɛ_{0}{\cos \left( {\zeta_{0} - \zeta_{R}} \right)}{\cos \left( {\phi_{R} + \zeta_{R}} \right)}} \\{{- ɛ_{0}}{\sin \left( {\zeta_{0} - \zeta_{R}} \right)}{\sin \left( {\phi_{R} + \zeta_{R}} \right)}}\end{matrix} \right\rbrack + \mspace{40mu} \left\lbrack \begin{matrix}{ɛ_{2}{\cos \left( {{{- 2}\zeta_{1}} + \zeta_{2} + \zeta_{R}} \right)}{\cos \left( {\phi_{R} + {2\phi} + {2\zeta_{1}} - \zeta_{R}} \right)}} \\{{- ɛ_{2}}{\sin \left( {{{- 2}\zeta_{1}} + \zeta_{2} + \zeta_{R}} \right)}{\sin \left( {\phi_{R} + {2\phi} + {2\zeta_{1}} - \zeta_{R}} \right)}}\end{matrix} \right\rbrack + \mspace{56mu} {\left\lbrack \begin{matrix}{ɛ_{3}{\cos \left( {{{- 3}\zeta_{1}} + \zeta_{3} + {2\zeta_{R}}} \right)}{\cos \left( {\phi_{R} + {3\phi} + {3\zeta_{1}} - {2\zeta_{R}}} \right)}} \\{{- ɛ_{3}}{\sin \left( {{{- 3}\zeta_{1}} + \zeta_{3} + {2\zeta_{R}}} \right)}{\sin \left( {\phi_{R} + {3\phi} + {3\zeta_{1}} - {2\zeta_{R}}} \right)}}\end{matrix} \right\rbrack.}}} & (65)\end{matrix}$

Equation (65) for S_(ψ) is next written in the form

$\begin{matrix}{{S_{\psi}(t)} = {\left\lbrack {{A_{- 1}{\cos \left( {\phi_{R} - \phi - \zeta_{1} + {2\zeta_{R}}} \right)}} - {B_{- 1}{\sin \left( {\phi_{R} - \phi - \zeta_{1} + {2\zeta_{R}}} \right)}}} \right\rbrack + \left\lbrack {{A_{0}{\cos \left( {\phi_{R} + \zeta_{R}} \right)}} - {B_{0}{\sin \left( {\phi_{R} + \zeta_{R}} \right)}}} \right\rbrack + \left\lbrack {{A_{2}{\cos \left( {\phi_{R} + {2\phi} + {2\zeta_{1}} - \zeta_{R}} \right)}} - {B_{2}{\sin \left( {\phi_{R} + {2\phi} + {2\zeta_{1}} - \zeta_{R}} \right)}}} \right\rbrack + \left\lbrack {{A_{3}{\cos \left( {\phi_{R} + {3\phi} + {3\zeta_{1}} - {2\zeta_{R}}} \right)}} - {B_{3}{\sin \left( {\phi_{R} + {3\phi} + {3\zeta_{1}} - {2\zeta_{R}}} \right)}}} \right\rbrack}} & (66)\end{matrix}$

where A⁻¹, B⁻¹, A₀, and B₀ are given by equations (37), (38), (22), and(23), respectively, and

A ₂=ε₂ cos(−2ζ₁+ζ₂+ζ_(R)),  (67)

B ₂=ε₂ sin(−2ζ₁+ζ₂+ζ_(R)),  (68)

A ₃=ε₃ cos(−3ζ₁+ζ₃+2ζ_(R)),  (69)

B ₃=ε₃ sin(−3ζ₁+ζ₃+2ζ_(R)).  (70)

Compensation error signal S_(ψ) is generated in processor 44 usingEquation (66), the coefficients transmitted from table 40 as signal 42,and the signals Σ⁻¹(t), {tilde over (Σ)}⁻¹(t), S_(R), {tilde over(S)}_(R), Σ₂(t), {tilde over (Σ)}₂(t), Σ₃(t), and {tilde over (Σ)}₃(t)(which comprise the cyclic error basis functions) under control ofcontrol of signal 74 (from processor 70). Explicitly,

$\begin{matrix}{{{S_{\psi}(t)} = {\left\lbrack {{\left( \frac{A_{R}^{2}A_{1}A_{- 1}}{\left( A_{R}^{2} \right)^{2}A_{1}^{2}} \right)\Sigma_{- 1}} + {\left( \frac{{- A_{R}^{2}}A_{1}B_{- 1}}{\left( A_{R}^{2} \right)^{2}A_{1}^{2}} \right){\overset{\sim}{\Sigma}}_{- 1}}} \right\rbrack + \left\lbrack {{\left( \frac{A_{R}A_{0}}{A_{R}^{2}} \right)S_{R}} - {\left( \frac{A_{R}B_{0}}{A_{R}^{2}} \right){\overset{\sim}{S}}_{R}}} \right\rbrack + \left\lbrack {{\left( \frac{A_{R}A_{1}^{2}A_{2}}{{A_{R}^{2}\left( A_{1}^{2} \right)}^{2}} \right)\Sigma_{2}} + {\left( \frac{{- A_{R}}A_{1}^{2}B_{2}}{{A_{R}^{2}\left( A_{1}^{2} \right)}^{2}} \right){\overset{\sim}{\Sigma}}_{2}}} \right\rbrack + \left\lbrack {{\left( \frac{A_{R}^{2}A_{1}^{3}A_{3}}{\left( A_{R}^{2} \right)^{2}\left( A_{1}^{2} \right)^{3}} \right)\Sigma_{3}} + {\left( \frac{{- A_{R}^{2}}A_{1}^{3}B_{3}}{\left( A_{R}^{2} \right)^{2}\left( A_{1}^{2} \right)^{3}} \right){\overset{\sim}{\Sigma}}_{3}}} \right\rbrack}}} & (71)\end{matrix}$

In other words, the compensation error signal is generated form asuperposition of the error basis functions weighted by the cyclic errorcoefficients.

The compensating signal S_(ψ) is subtracted from signal S in processor46 under control of signal 76 (from processor 70) to generatecompensated signal S−S_(ψ). Control signal 76 determines when signal Sis to be compensated. The phase φ=4kL is then extracted from thecompensated signal with a subsequent processor (not shown) to, forexample, provide a more accurate measurement of the distance L.

In the presently preferred embodiment, the error compensation signalS_(ψ)(t) is subtracted from prior values of the signals S(t). Forexample, feedforward signal S′(t), (as described in Equation (14)) mayreplace signal S(t). The delay, m, of the feedforward signal is chosento be equal to the processing delay in calculating the error basisfunctions and subsequently S_(ψ)(t) from signal S(t). In this manner,S′(t) and S_(ψ)(t) represent the same time of input signal S(t).

In further embodiments, the cyclic error coefficients may be stored andupdated at a lower data rate than that used to generate the cyclic errorbasis functions from the feedforward values. In such cases, the storedvalues for the cyclic error coefficients may used for the calculation ofthe cyclic error basis functions as necessary. Of course, in yet furtherembodiments, the coefficients and/or the error basis functions can becalculated in real time, without the use of the feedforward signals.

An important assumption in the preferred embodiment of the cyclic errorcompensation described above is that the Doppler shift frequencydφ(t)/dt is small enough relative to the heterodyne frequency ω_(R),that the quadrature signal S(t) can be approximated (in analogy toEquation (10) by the expression:

$\begin{matrix}{{\overset{\sim}{S}(t)} = {{\left( {\cot \; \omega_{R}\tau} \right){S\left( {t - {2\tau}} \right)}} - {\frac{\cos \; 2\; \omega_{R}\tau}{\sin \; \omega_{R}\tau}{S\left( {t - \tau} \right)}}}} & (72)\end{matrix}$

or the simpler expression given by Equation (12). In furtherembodiments, the quadrature signal {tilde over (S)}(t) may be moreaccurately calculated according to:

$\begin{matrix}{{\overset{\sim}{S}(t)} = {{\left( {\cot \; \omega_{M}\tau} \right){S\left( {t - {2\tau}} \right)}} - {\frac{\cos \; 2\; \omega_{M}\tau}{\sin \; \omega_{M}\tau}{S\left( {t - \tau} \right)}}}} & (73)\end{matrix}$

where ω_(M) is the instantaneous rate of change of the phase of the maininterference signal S(t), which can be determined with sufficientaccuracy by assuming that the cyclic error contributions to S(t) arenegligible.

Also, in further embodiments, the cyclic error compensation techniquemay be used for cyclic error terms different from those explicitlydescribed in Equations (3)-(6). In particular, using algebraiccombinations of the signals S(t), {tilde over (S)}(t), S_(R)(t), and{tilde over (S)}_(R)(t), a processing unit can generate cyclic errorbasis functions, which are sine and cosine functions that have the sametime-varying arguments as the cyclic error terms that need to becompensated, and then use the cyclic error basis functions to projectout respective cyclic error coefficients from S(t) and {tilde over(S)}(t) by low-pas filtering (e.g., averaging).

For example, to determine the coefficients for a half-cycle cyclic errorof the form:

$\begin{matrix}{S_{\psi {({1/2})}} = {ɛ_{1/2}{\cos \left( {\phi_{R} + \frac{\phi}{2} + \zeta_{1/2}} \right)}}} & (74)\end{matrix}$

one can calculate cyclic error basis functions for the half-cycle cyclicerror as follows. First calculate signals Σ′₇(t) and {tilde over(Σ)}′₇(t) as:

$\begin{matrix}{{{\Sigma_{7}^{\prime}(t)} = \sqrt{\frac{\Sigma_{0} - \left( {{A_{R}A_{0}} - {A_{R}A_{1}}} \right)}{2A_{R}A_{1}}}},} & (75) \\{{{\overset{\sim}{\Sigma}}_{7}^{\prime}(t)} = {\sqrt{\frac{\left( {{A_{R}A_{0}} + {A_{R}A_{1}}} \right) - \Sigma_{0}}{2A_{R}A_{1}}}.}} & (76)\end{matrix}$

Notably, the leading term of Σ′₇(t) is |cos(φ/2+ζ₁/2−ζ_(R)/2)|, and theleading term of {tilde over (Σ)}′₇(t) is |sin(φ/2+ζ₁/2−ζ_(R)/2)|. Zerophase crossings in Σ′₇(t) and {tilde over (Σ)}′₇(t) are then measured toremove the absolute value operation and define Σ₇(t) and {tilde over(Σ)}₇(t), which have leading terms cos(φ/2+ζ₁/2−ζ_(R)/2) andsin(φ/2+ζ₁/2−ζ_(R)/2), respectively.

Half-cycle error basis functions Σ_(1/2)(t) and {tilde over(Σ)}_(1/2)(t) are then generated as:

Σ_(1/2)(t)=Σ₇(t)S _(R)(t)−{tilde over (Σ)}₇(t){tilde over (S)}_(R)(t),  (77)

{tilde over (Σ)}_(1/2)(t)={tilde over (Σ)}₇(t)S _(R)(t)+Σ₇(t){tilde over(S)} _(R)(t),  (78)

where Σ_(1/2) (t) and {tilde over (Σ)}_(1/2) (t) have as their leadingterms sine and cosine functions, respectively, whose time-dependentargument is the same as that of S_(ψ(1/2)). To determine thecoefficients for S_(ψ(1/2)), one projects half-cycle error basisfunctions onto S(t) and {tilde over (S)}(t) to move the half-cyclecyclic error component to zero frequency, for example:

Σ₈(t)=S(t)Σ_(1/2)(t)+{tilde over (S)}(t){tilde over (Σ)}_(1/2)(t),  (79)

{tilde over (Σ)}₈(t)={tilde over (S)}(t)Σ_(1/2)(t)−S(t){tilde over(Σ)}_(1/2)(t).  (80).

Low-pass filtering (e.g., with the Butterworth filter) of Σ₈(t) and{tilde over (Σ)}₈(t) then yield the half-cycle error coefficients inanalogy to the extraction of the previously described cyclic errorcoefficients. In particular, the leading terms following the low-passfiltering are A_(R)A_(1/2) cos(ζ_(1/2)−ζ₁/2−ζ_(R)/2) and A_(R)A_(1/2)sin(ζ_(1/2)−ζ₁/2−ζ_(R)/2), respectively.

We now describe the CMS approach for the CEC, in which a cyclic errorcorrection signal D_(ψ)(t) is subtracted from a corresponding DFT signalD(t) of the electrical interference signal S(t) to produce a compensatedDFT signal. FIG. 4 c shows a simplified schematic diagram of ameasurement using the CMS approach. The optical interference signal 111is received and amplified by photoelectric receiver 112. The resultingelectrical interference signal 113 is filtered by lowpass filter (LPF)114 producing filtered signal 115. The LPF 114 is designed to preventharmonics of the interference signal 111 from being aliased into thefrequency range of interest. Filtered signal 115 is digitized by ADC116, to produce digitized measurement signal 117. A typical ADC for ahigh performance displacement measuring interferometer may have 12 bitsof resolution at sampling rates of 120 MHz. The digitized measurementsignal 117 is processed by phase meter 120 (described below) to produceoutputs magnitude 125 and phase 127 which represent the digitizedmeasurement signal 117 as a transform. The magnitude output 125 is usedfor status and diagnostic purposes. The phase output 127 is used byposition calculator 130 which is fully described in published U.S.application Ser. No. 10/211,435 (publication number US 2003/0025914 A1),incorporated herein by reference. Position calculator 130 calculatesmeasured position 131 and estimated speed 133. The measured position 131is filtered by digital filter 136, which is fully described in U.S. Pat.No. 5,767,972, incorporated herein by reference, to generate filteredposition signal 137. Filtered position signal 137 represents the desiredmeasurement of the distance L.

Phase meter 120 includes a Discrete Fourier Transform (DFT) processor122, a cyclic error compensation (CEC) calculator 140, and a CoordinateRotation by Digital Computer (CORDIC) converter 124. Signals 123, 143,145, and 147 are complex values, which consist of both a real componentand an imaginary component, as a+jb, where a is the real component, b isthe imaginary component, and j is √{square root over (−1)}. (The symboli is sometimes used in the literature instead of j.) Otherrepresentations of complex or quadrature values can be used, and may beexpressed using other symbols such as, for example, I and Q, or X and Y,or A and Ã. Complex values may be converted from rectangular (real andimaginary) representation to polar (magnitude and phase angle)representation. The numeric representation of the digital signals may beinteger, fractional, or floating point.

The DFT processor 122 converts a series of consecutive samples ofdigitized measurement signal 117 into a complex measurement signal 123representing a transform of the digitized measurement signal 117 at aselected center frequency of DFT processor 122. The center frequency isdetermined by control circuitry (not shown) and the estimated speed 133is determined by position calculator 130.

An exemplary DFT processor 122 is a 72-point windowed DFT performed at a10 MHz rate. Since the DFT calculation is being updated in time, thecomplex measurement signal 123 is expressed as a function of time D(t).This 10 MHz update rate provides 83% overlap of samples of the digitizedmeasurement signal 117 between one DFT calculation and the next for anADC sampling rate of f_(S)=120 MHz. A typical window function is theBlackman window, which reduces errors due to the discontinuities at thebeginning and end of the series of digitized measurement signal samplesused for the DFT.

The CEC calculator 140 calculates and compensates for certain of thecyclic errors. CEC error estimator 144 (described in more detail belowwith reference to FIG. 4 d) calculates complex error compensation signal145. Optional delay 142, and other delays (not shown) in CEC calculator140 may be used to match the processing delay of the variouscalculations. Adder 146 combines delayed complex measurement signal 143with complex error compensation signal 145 to produce compensatedcomplex measurement signal 147, in which the certain cyclic errorsignals are substantially reduced.

CORDIC converter 124 converts the compensated complex measurement signal147 to magnitude 125 and phase 127.

Referring to FIG. 4 d, the CEC error estimator 144 includes twoprocessing units. One processing unit 148 determines error basisfunctions and complex factors relating to the amplitudes and offsetphases of the certain cyclic errors that need be compensated. A secondprocessing unit 204 generates complex error compensation signal D_(ψ)(t)145 using the error basis functions and complex factors relating to theamplitudes and offset phases determined by first processing unit 148.

The first processing unit 148 for one embodiment is shown schematicallyin FIG. 4 d and the second processing unit 204 of this embodiment isshown schematically in FIG. 4 e. These processing units are incorporatedinto the architecture shown in FIG. 4 c that may also include any of avariety of other techniques such as a glitch filter (as described inpublished U.S. application Ser. No. 10/211,435), dynamic data ageadjustment (as described in U.S. Pat. No. 6,597,459, incorporated hereinby reference), and digital filtering as described in U.S. Pat. No.5,767,972.

Note that, as in the DMS approach, the set of cyclic error terms whosephases are independent of φ_(R) has been omitted from Equation (2)because, in this CMS approach, they are eliminated by LPF (Lowpassfilter) 114 and the bandpass filtering inherent in DFT processor 122.

Note that there are conditions where the cyclic error terms 82 or 83 maybe aliased and appear at or near the frequency of the measurementsignal. Let f_(R) be the reference frequency (f_(R)=(O_(R)/2π), f_(D) bethe Doppler shift frequency (f_(D)=ω_(D) /2π), and f_(M) be theresulting measurement frequency. When f_(D)=−(⅔)f_(R),

$\begin{matrix}{{f_{M} = {{f_{R} + f_{D}} = {{f_{R} - {\frac{2}{3}f_{R}}} = {\frac{1}{3}f_{R}}}}},{f_{ɛ_{2}} = {{f_{R} + {2f_{D}}} = {{f_{R} - {\frac{4}{3}f_{R}}} = {{- \frac{1}{3}}{f_{R}.{Also}}}}}},{{{when}\mspace{14mu} f_{D}} = {{- \left( {1/2} \right)}f_{R}}},} & (81) \\{{f_{M} = {{f_{R} + f_{D}} = {{f_{R} - {\frac{1}{2}f_{R}}} = {\frac{1}{2}f_{R}}}}},{f_{ɛ_{3}} = {{f_{R} + {3f_{D}}} = {{f_{R} - {\frac{3}{2}f_{R}}} = {{- \frac{1}{2}}{f_{R}.}}}}}} & (82)\end{matrix}$

In both cases, the negative frequency is indistinguishable from thecorresponding positive frequency, and measurement accuracy is affected.

Complex factors relating to amplitudes ε_(p) and offset phases ζ_(p) ofthe four cyclic error terms and time dependent factors of the cyclicerror terms are generated using processed values D(t) 123 from DFTprocessor 122. The factors are stored in registers 162, 176, 186, and192 for subsequent use in generation of the cyclic error correctionsignal D_(ψ)(t) 145. The time dependent factors of the cyclic errorterms are obtained by application of digital transforms based ontrigonometric identities and properties of complex signals.

DFT processor 122 calculates the complex DFT of the digitizedmeasurement signal 117 as:

$\begin{matrix}{{{D_{q}\left( t_{1} \right)} = {\sum\limits_{n = 0}^{N - 1}{W_{n}{S\left( {{n\; \tau} + t_{1}} \right)}\; \exp \; \left\{ {- {j\left( {2{\pi \left( {n - \frac{N - 1}{2}} \right)}{q/N}} \right)}} \right\}}}},} & (83)\end{matrix}$

for 0<q<N−1, where τ=1/f_(S), W_(n) is a window function centered at

${n = \frac{N - 1}{2}},$

and t₁ is the time at which the DFT calculation is updated. When q isselected by control circuitry (not shown) as an integer approximatelyequal to Nf_(M)/f_(S), to correspond to the center frequency of theprimary component of the digitized measurement signal. A typical valuefor N is 72, and a typical window function W_(n) is the Blackman windowfunction.

The equation for the DFT is arranged so that the center of the sequenceof the N consecutive samples of the digitized measurement signal 117(n=35.5 in this case) corresponds to zero phase and the center of thewindow function. Therefore, since the window function and DFTcoefficients (the complex exponential or equivalently the “cos+j sin”factors shown below) are symmetrical about the center, and windowweighting is zero at the ends, a change in the value of q, has a reducedeffect on the phase of the result.

The transform signal D_(q)(t₁) is updated at a rate f_(U) that is lowerthan the rate f_(S) at which the signal S(t) is sampled. In thisexample, t₁=lΔt₁ (where l is an integer and Δt₁≡1/f_(U) is the updateinterval) and f_(U)=10 MHz.

Alternatively, the DFT equation can be “folded” to reduce the number ofmultiplication operations that are performed and calculated as:

$\begin{matrix}{{D_{q}\left( t_{1} \right)} = {\sum\limits_{n = 0}^{\frac{N}{2} - 1}{{W_{n}\left( {{x_{n}\left( t_{1} \right)} + {x_{N - 1 - n}\left( t_{1} \right)}} \right)}\begin{pmatrix}{{\cos \left( {2\pi \frac{n + 0.5}{N}\frac{q}{8}} \right)} +} \\{j\; \sin \; \left( {2\pi \frac{n + 0.5}{N}\frac{q}{8}} \right)}\end{pmatrix}}}} & (84)\end{matrix}$

for 0<q<8N−1, where x_(n)(t₁) for n=0, . . . , N−1 are N consecutivesamples (i.e., t=nτ) of S(t+t₁), and W_(n) is centered at n=0. In thisexample, the larger range of q yields a more finely spaced resolution of⅛ bin to reduce amplitude variations (or “picket fence” effect) as thefrequency changes from one bin to the next.

The DFT function is equivalent to a mixing and a filtering operation.The mixing is a result of multiplying the input data by the complexexponential or equivalently the “cos+j sin” factor. The filtering is aresult of the summation and the window function W_(n).

Since the window function W_(n) makes terms zero outside the range ofsummation, the DFT expression can be written as a sum over all n. In asimplified expression for an “unfolded” DFT (as in equation (83)), thedigitized measurement signal S(nτ+t₁) 177 has been approximated by itsprimary term A₁ cos(φ_(R)(nτ+t₁)+φ(nτ+t₁)+ζ₁), where explicit sampling(t=nτ) has been included:

$\begin{matrix}{{D\left( t_{1} \right)} = {A_{1}{\sum\limits_{n}{W_{n}{\cos \begin{pmatrix}{{\phi_{R}\left( {{n\; \tau} + t_{1}} \right)} +} \\{{\phi \left( {{n\; \tau} + t_{1}} \right)} + \zeta_{1}}\end{pmatrix}}\begin{pmatrix}{{\cos \left( {\phi_{C}\left( {n,q} \right)} \right)} +} \\{{jsin}\left( {\varphi_{C}\left( {n,q} \right)} \right)}\end{pmatrix}}}}} & (85)\end{matrix}$

where (φ_(R)(nτ+t₁)=ω_(R) nτ+ω_(R)t₁ and φ(nτ+t₁)=ω_(D)nτ+ω_(D)t₁+φ₀ isthe sampled version of the Doppler phase shift φ(t)=ω_(D)t+φ₀, and

${\phi_{C}\left( {n,q} \right)} = {2\pi \; \frac{nq}{N}}$

is the phase of the DFT with q≈Nf_(M)/f_(S), yieldingφ_(C)(n,q)≈2πf_(M)nτ=ω_(M)nτ. (The q in D_(q)(t₁) has been suppressed sothat D(t₁)≡D_(q)(t₁).) Using trigonometric identities, equation (85) canbe expanded to (with time dependent arguments temporarily suppressed):

$\begin{matrix}{{D\left( t_{1} \right)} = {A_{1}{W_{n}\begin{bmatrix}{{\frac{1}{2}\begin{pmatrix}{{\cos \left( {\phi_{R} + \phi + \zeta_{1} + \phi_{C}} \right)} +} \\{\cos \left( {\phi_{R} + \phi + \zeta_{1} - \phi_{C}} \right)}\end{pmatrix}} +} \\{j\frac{1}{2}\begin{pmatrix}{{\sin \left( {\phi_{R} + \phi + \zeta_{1} + \phi_{C}} \right)} -} \\{\sin \left( {\phi_{R} + \phi + \zeta_{1} - \phi_{C}} \right)}\end{pmatrix}}\end{bmatrix}}}} & (86)\end{matrix}$

The terms containing φ_(R)+φ+ζ₁+φ_(C) are high frequency sinusoidsvarying with n that are filtered out in the summation including thewindow function W_(n) that covers many cycles. The constant ½ may bedropped for convenience. The terms containing (φ_(R)+φ+ζ₁−φ_(C) that areslowly varying in the summation over the window remain:

$\begin{matrix}{{D\left( t_{1} \right)} \approx {A_{1}{\sum\limits_{n}{W_{n}\begin{bmatrix}{{\cos \begin{pmatrix}{{\omega_{R}n\; \tau} + {\omega_{R}t_{1}} + {\omega_{D}n\; \tau} +} \\{{\omega_{D}t_{1}} + \phi_{0} + \zeta_{1} - {\omega_{M}n\; \tau}}\end{pmatrix}} -} \\{j\; {\sin \begin{pmatrix}{{\omega_{R}n\; \tau} + {\omega_{R}t_{1}} + {\omega_{D}n\; \tau} +} \\{{\omega_{D}t_{1}} + \phi_{0} + \zeta_{1} - {\omega_{M}n\; \tau}}\end{pmatrix}}}\end{bmatrix}}}}} & (87)\end{matrix}$

Since ω_(M)≈ω_(R)+ω_(D) the terms of the sinusoids including n drop out.

When the update interval Δt₁≡1/f_(U) is given by Δt₁=m/f_(R) (where m isan integer) the value of t₁ is a multiple of 1/f_(R). Under thiscondition, for low stage velocities (i.e., ω_(D)<<ω_(R))ω_(R)t₁≈m2π andDFT calculation becomes:

$\begin{matrix}\begin{matrix}{{D\left( t_{1} \right)} \approx {A_{1}{\sum\limits_{n}{W_{n}\begin{bmatrix}{{\cos \left( {{\omega_{D}t_{1}} + \phi_{0} + \zeta_{1}} \right)} -} \\{j\; {\sin \left( {{\omega_{D}t_{1}} + \phi_{0} + \zeta_{1}} \right)}}\end{bmatrix}}}}} \\{\approx {A_{1}\left\lbrack {{\cos \left( {{\omega_{D}t_{1}} + \phi_{0} + \zeta_{1}} \right)} - {j\; {\sin \left( {{\omega_{D}t_{1}} + \phi_{0} + \zeta_{1}} \right)}}} \right\rbrack}}\end{matrix} & (89)\end{matrix}$

The DFT result is a complex value which can represent either a positiveor a negative frequency. Equation (88) could be written as a sum, ratherthan a difference, with appropriate changes in sign in other equations.This choice of sign is not important, provided that the choices resultin a reduction in cyclic error.

Equation (88) can be combined with equations (1) to (6). This resultsin:

D(t ₁)=A ₁(cos(φ(t ₁)+ζ₁)−j sin(φ(t ₁)+ζ₁))+D _(ψ)(t ₁)  (89)

where

D _(ψ)(t ₁)=D _(ψ−1)(t ₁)+D _(ψ0) +D _(ψ2)(t ₁)+D _(ψ3)(t ₁);  (90)

D _(ψ−1)(t ₁)=ε⁻¹(cos(−φ(t ₁)+ζ⁻¹)−j sin(−φ(t ₁)+ζ⁻¹)),  (91)

D _(ψ0)=ε₀(cos(ζ₀)−j sin(ζ₀)),  (92)

D _(ψ2)(t ₁)=ε₂(cos(2φ(t ₁)+₂)−j sin(2φ(t ₁)+ζ₂)),  (93)

D _(ψ3)(t ₁)=ε₃(cos(3φ(t ₁)+ζ₃)−j sin(3φ(t ₁)+Λ₃));  (94)

where φ(t₁) is the value of the Doppler phase shift φ(t₁)=ω_(D)t₁+φ₀that is updated with t₁. For convenience, in the equations (98)-(116)below, t₁ is written simply as t and φ(t₁) is written simply as φ. Also,complex measurement signal D(t) 123 and complex error compensationsignal D_(ψ)(t) 145 are assumed to be updated at the rate f_(U) suchthat D(t)≡D(t₁) and D_(ψ)(t)≡D_(ψ)(t₁).

FIG. 4 d shows a schematic diagram of CEC error estimator 144. In thefollowing discussions the product of two complex signals frequentlyoccurs. The following identities are used:

(cos(A)+j sin(A))·(cos(B)+j sin(B))=cos(A+B)+j sin(A+B),  (95)

(cos(A)+j sin(A))·(cos(B)−j sin(B))=cos(A−B)+j sin(A−B),  (96)

(cos(A)+j sin(A))*=cos(A)−j sin(A)=cos(−A)+j sin(−A)  (97)

The next step is the processing of signals for information about thecyclic error terms. In a first “lowpass filtering approach,” the signalD(t) is sent to LPF (Lowpass Filter) 160, for example an IIR (InfiniteImpulse Response) Butterworth digital filter, an FIR (Finite ImpulseResponse), or CIC (Cascaded Integrator Comb) digital filter as describedby Hogenauer (An Economical Class of Digital Filters for Decimation andInterpolation; E. B. Hogenauer; IEEE Transactions on Acoustics, Speech,and Signal Processing; Vol ASSP-29, No 2, April 1981, p 155-162,incorporated herein by reference). The CIC filter has the advantages inthis implementation of simple design (using only integer addition) anddecimation by large ratios. The implementation of an LPF for a complexsignal uses two identical real LPF functions, one is used for the realcomponent, and one is used for the imaginary component. The use ofdigital functions ensures precise matching of amplitude and phaseresponse of the two filters.

For an LPF T_(n)(x) of order n, the complex output C₀ is approximately:

$\begin{matrix}\begin{matrix}{C_{0} = {T_{n}\left\lbrack {D(t)} \right\rbrack}} \\{= {{ɛ_{0}\left( {{\cos \; \left( \zeta_{0} \right)} - {j\; {\sin \left( \zeta_{0} \right)}}} \right)} +}} \\{\left\lbrack {{A_{1}{O\left( \frac{\omega_{c}}{\omega_{D}} \right)}^{n}} + {ɛ_{- 1}{O\left( \frac{\omega_{c}}{\omega_{D}} \right)}^{n}} + {ɛ_{2}{O\left( \frac{\omega_{c}}{2\omega_{D}} \right)}^{n}} + {ɛ_{3}{O\left( \frac{\omega_{c}}{3\omega_{D}} \right)}^{n}}} \right\rbrack}\end{matrix} & (98)\end{matrix}$

where O(x) denotes a term of the order of x, ω_(c) is the −3 dB angularcutoff frequency, and ω_(D)=dφ/dt.

The term on the right hand sides of Equation (98) with factor A₁ is thesource of the largest error and accordingly determines thespecifications of n and the minimum ratio for ω_(D)/ω_(c) that can beused when the outputs of LPF 160 are stored in register 162. For afourth order LPF, i.e., n=4, and a minimum ratio for ω_(D)/ω_(c)=7, theerror terms on the right hand side of Equation (98) will generate errorsthat correspond to ≲0.010 nm (3σ). When the stage is moving at a speedsuch that the corresponding Doppler shift frequency ω_(D) /2π is 10 to100 times greater than the bandwidth of the stage servo control systemand the requirement with respect to ω_(D)/ω_(c) is satisfied, the outputC₀ of the LPF 160 is stored in register 162 as CO_(R) under the controlof signal 161. This stage speed requirement reduces the possibility thatsidebands of the primary Doppler signal caused by actual variations inthe stage position or motion will be interpreted as cyclic errors.

An advantage of this approach is that ω_(D) can vary by factors such as2 or more during the period when output values of C₀ are stored inregister 162.

In a second approach to processing signals for information about thecyclic error terms, the CEC error estimator 144 stores values in theregisters based on analysis of collective properties of a distributionof values of D(t). An advantage of this approach is that the stage canbe nearly stationary, or moving at a speed such that the correspondingDoppler shift frequency ω_(D) /2π is less than 10 times greater than thebandwidth of the stage servo control system. In this case, the measuredmotion typically has negligible sidebands that could be interpreted ascyclic errors. This “distribution analysis approach” is described inmore detail below.

The values for C_(0R) and D(t) are transmitted to processor 164 for thegeneration of complex signal Σ₁(t) where

$\begin{matrix}\begin{matrix}{{\Sigma_{1}(t)} = {{D(t)} - C_{0\; R}}} \\{= {{A_{1}\left( {{\cos \left( {\phi + \zeta_{1}} \right)} - {j\; \sin \; \left( {\phi + \zeta_{- 1}} \right)}} \right)} +}} \\{{{ɛ_{- 1}\left( {{\cos \left( {{- \phi} + \zeta_{- 1}} \right)} - {{jsin}\; \left( {{- \phi} + \zeta_{- 1}} \right)}} \right)} +}} \\{{{ɛ_{2}\left( {{\cos \left( {{2\phi} + \zeta_{2}} \right)} - {{jsin}\; \left( {{2\phi} + \zeta_{2}} \right)}} \right)} +}} \\{{{ɛ_{3}\left( {{\cos \left( {{3\phi} + \zeta_{3}} \right)} - {{jsin}\; \left( {{3\phi} + \zeta_{3}} \right)}} \right)}.}}\end{matrix} & (99)\end{matrix}$

Signal Σ₁ is sent to processor 168, which calculates Σ⁻¹ as the complexconjugate of Σ₁.

$\begin{matrix}\begin{matrix}{{\Sigma_{- 1}(t)} = {\Sigma_{1}(t)}^{*}} \\{= {{A_{1}\left( {{\cos \left( {{- \phi} - \zeta_{1}} \right)} - {j\; \sin \; \left( {{- \phi} - \zeta_{1}} \right)}} \right)} +}} \\{{{ɛ_{- 1}\left( {{\cos \left( {\phi - \zeta_{- 1}} \right)} - {{jsin}\; \left( {\phi - \zeta_{- 1}} \right)}} \right)} +}} \\{{{ɛ_{2}\left( {{\cos \left( {{{- 2}\phi} - \zeta_{2}} \right)} - {{jsin}\; \left( {{{- 2}\phi} - \zeta_{2}} \right)}} \right)} +}} \\{{{ɛ_{3}\left( {{\cos \left( {{{- 3}\phi} - \zeta_{3}} \right)} - {{jsin}\; \left( {{{- 3}\phi} - \zeta_{3}} \right)}} \right)}.}}\end{matrix} & (100)\end{matrix}$

Signal Σ₁ is sent to processor 180, which calculates Σ₂.

$\begin{matrix}\begin{matrix}{{\Sigma_{2}(t)} = \left( \Sigma_{1} \right)^{2}} \\{= {{A_{1}^{2}\left( {{\cos \left( {{2\phi} + {2\zeta_{1}}} \right)} - {j\; \sin \; \left( {{2\phi} + {2\zeta_{1}}} \right)}} \right)} +}} \\{{{2A_{1}{ɛ_{- 1}\left( {{\cos \left( {\zeta_{1} + \zeta_{- 1}} \right)} - {{jsin}\; \left( {\zeta_{1} + \zeta_{- 1}} \right)}} \right)}} +}} \\{{{2A_{1}{ɛ_{2}\left( {{\cos \left( {{3\phi} + \zeta_{1} + \zeta_{2}} \right)} - {{jsin}\; \left( {{3\phi} + \zeta_{1} + \zeta_{2}} \right)}} \right)}} +}} \\{{{2A_{1}{ɛ_{3}\left( {{\cos \left( {{4\phi} + \zeta_{1} + \zeta_{3}} \right)} - {{jsin}\; \left( {{4\phi} + \zeta_{1} + \zeta_{3}} \right)}} \right)}} +}} \\{{{O\left( {ɛ_{i},ɛ_{j}} \right)}.}}\end{matrix} & (101)\end{matrix}$

Signal Σ₂ is divided by two and sent to LPF 190, as described earlierfor LP 160.

For an LPF T_(n)(x) of order n, the complex output C₄ is approximately:

$\begin{matrix}\begin{matrix}{C_{4} = {T_{n}\left\lbrack {\frac{1}{2}\Sigma_{2}} \right\rbrack}} \\{= {{A_{1}{ɛ_{- 1}\left( {{\cos \left( {\zeta_{1} + \zeta_{- 1}} \right)} - {{jsin}\left( {\zeta_{1} + \zeta_{- 1}} \right)}} \right)}} +}} \\{\left\lbrack {{\frac{1}{2}A_{1}^{2}{O\left( \frac{\omega_{c}}{2\omega_{D}} \right)}^{n}} + {A_{1}ɛ_{2}{O\left( \frac{\omega_{c}}{3\omega_{D}} \right)}^{n}} + {A_{1}ɛ_{3}{O\left( \frac{\omega_{c}}{4\omega_{D}} \right)}^{n}}} \right\rbrack}\end{matrix} & (102)\end{matrix}$

where O(x) denotes a term of the order of x, ω_(c) is the −3 dB angularcutoff frequency, and ω_(D)=dφ/dt.

The term on the right hand sides of Equation (102) with factors A₁ ² isthe source of the largest error and accordingly determines thespecifications of n and the minimum ratio for ω_(D)/ω_(c) that can beused when the outputs of processor 190 are stored in register 192. For afourth order LPF, i.e., n=4, and a minimum ratio for ω_(D)/ω_(c)=3.5,the error terms on the right hand side of Equation (102) will generateerrors that correspond to ≲0.010 nm (3σ). When the stage is moving at aspeed such that the corresponding Doppler shift frequency ω_(D) /2π is10 to 100 times greater than the bandwidth of the stage servo controlsystem and the requirement with respect to ω_(D)/ω_(c) is satisfied, theoutput C₄ of LPF 190 is stored in register 192 as C_(4R) under thecontrol of signal 161.

Signals Σ₁, Σ₂, and C_(4R) are sent to processor 200 which calculatesΣ₃. Signal Σ_(3A) is calculated by combining signals C_(4R) and Σ₁ usingmultiplier 194:

$\begin{matrix}\begin{matrix}{\Sigma_{3A} = {C_{4R}\Sigma_{1}}} \\{= {{A_{1}^{2}{ɛ_{- 1}\left( {{\cos \left( {{2\zeta_{1}} + \zeta_{- 1} + \phi} \right)} - {{jsin}\left( {{2\zeta_{1}} + \zeta_{- 1} + \phi} \right)}} \right)}} +}} \\{{{A_{1}{ɛ_{- 1}^{2}\left( {{\cos \left( {\zeta_{1} + {2\zeta_{- 1}} - \phi} \right)} - {{jsin}\left( {\zeta_{1} + {2\zeta_{- 1}} + \phi} \right)}} \right)}} +}} \\{{A_{1}ɛ_{- 1}{ɛ_{2}\left( {{\cos \left( {\zeta_{1} + \zeta_{- 1} + {2\phi} + \zeta_{2}} \right)} -} \right.}}} \\{\left. {{jsin}\left( {\zeta_{1} + \zeta_{- 1} + {2\phi} + \zeta_{2}} \right)} \right) +} \\{{A_{1}ɛ_{- 1}{ɛ_{3}\left( {{\cos \left( {\zeta_{1} + \zeta_{- 1} + {3\phi} + \zeta_{3}} \right)} -} \right.}}} \\{\left. {{jsin}\left( {\zeta_{1} + \zeta_{- 1} + {3\phi} + \zeta_{3}} \right)} \right) + {A_{1}ɛ_{- 1}{{O\left( {ɛ_{i},ɛ_{j}} \right)}.}}}\end{matrix} & (103)\end{matrix}$

Signal Σ_(3B) is calculated by combining signals Σ₁ and Σ₂ usingmultiplier 202:

$\begin{matrix}\begin{matrix}{\Sigma_{3B} = {\Sigma_{1}\Sigma_{2}}} \\{= {{A_{1}^{3}\left( {{\cos \left( {{3\phi} + {3\zeta_{1}}} \right)} - {{jsin}\left( {{3\phi} + {3\zeta_{1}}} \right)}} \right)} +}} \\{{{3A_{1}^{2}{ɛ_{- 1}\left( {{\cos \left( {\phi + {2\zeta_{1}} + \zeta_{- 1}} \right)} - {{jsin}\left( {\phi + {2\zeta_{1}} + \zeta_{- 1}} \right)}} \right)}} +}} \\{{{3A_{1}^{2}{ɛ_{2}\left( {{\cos \left( {{4\phi} + {2\zeta_{1}} + \zeta_{2}} \right)} - {{jsin}\left( {{4\phi} + {2\zeta_{1}} + \zeta_{2}} \right)}} \right)}} +}} \\{{{3A_{1}^{2}{ɛ_{3}\left( {{\cos \left( {{5\phi} + {2\zeta_{1}} + \zeta_{3}} \right)} - {{jsin}\left( {{5\phi} + {2\zeta_{1}} + \zeta_{3}} \right)}} \right)}} +}} \\{{{O\left( {ɛ_{i},ɛ_{j}} \right)}.}}\end{matrix} & (104)\end{matrix}$

Signal Σ₃ is calculated by combining signals Σ_(3A) and Σ_(3B) usingconstant multiplier 196 and subtractor 198:

$\begin{matrix}\begin{matrix}{\Sigma_{3} = {\Sigma_{3B} - {3\Sigma_{3A}}}} \\{= {{A_{1}^{3}\left( {{\cos \left( {{3\phi} + {3\zeta_{1}}} \right)} - {{jsin}\left( {{3\phi} + {3\zeta_{1}}} \right)}} \right)} +}} \\{{{3A_{1}^{2}{ɛ_{2}\left( {{\cos \left( {{4\phi} + {2\zeta_{1}} + \zeta_{2}} \right)} - {{jsin}\left( {{4\phi} + {2\zeta_{1}} + \zeta_{2}} \right)}} \right)}} +}} \\{{{3A_{1}^{2}{ɛ_{3}\left( {{\cos \left( {{5\phi} + {2\zeta_{1}} + \zeta_{3}} \right)} - {{jsin}\left( {{5\phi} + {2\zeta_{1}} + \zeta_{3}} \right)}} \right)}} +}} \\{{3A_{1}{{O\left( {ɛ_{i},ɛ_{j}} \right)}.}}}\end{matrix} & (105)\end{matrix}$

Signal 5 is calculated by combining signals Σ⁻¹ and Σ₂ using multiplier182:

$\begin{matrix}\begin{matrix}{\Sigma_{5} = {\Sigma_{- 1}\Sigma_{2}}} \\{= {{A_{1}^{2}{ɛ_{2}\left( {{\cos \left( {{2\zeta_{1}} - \xi_{2}} \right)} - {{jsin}\left( {{2\zeta_{1}} - \zeta_{2}} \right)}} \right)}} +}} \\{{{A_{1}^{2}\begin{Bmatrix}{{+ {ɛ_{1}\begin{pmatrix}{{2\left( {{\cos \left( {{- \phi} + \zeta_{- 1}} \right)} - {{jsin}\left( {{- \phi} - \zeta_{1}} \right)}} \right)} +} \\\left( {{\cos \left( {{3\phi} + {2\zeta_{1}} - \zeta_{1}} \right)} -} \right. \\\left. {{jsin}\left( {{3\phi} + {2\zeta_{1}} - \zeta_{- 1}} \right)} \right)\end{pmatrix}}} +} \\{{A_{1}\left( {{\cos \left( {\phi + \zeta_{1}} \right)} - {{jsin}\left( {\phi + \zeta_{1}} \right)}} \right)} +} \\{{2{ɛ_{2}\left( {{\cos \left( {{2\phi} + \zeta_{2}} \right)} - {{jsin}\left( {{2\phi} + \zeta_{2}} \right)}} \right)}} +} \\{ɛ_{3}\begin{pmatrix}{{2\left( {{\cos \left( {{3\phi} + \zeta_{3}} \right)} - {{jsin}\left( {{3\phi} + \zeta_{3}} \right)}} \right)} +} \\\left( {{\cos \left( {{- \phi} + {2\zeta_{1}} - \zeta_{3}} \right)} - {{jsin}\left( {{- \phi} + {2\zeta_{1}} - \zeta_{3}} \right)}} \right)\end{pmatrix}}\end{Bmatrix}} +}} \\{{{A_{1}{O\left( ɛ_{i} \right)}} + {\ldots \mspace{11mu}.}}}\end{matrix} & (106)\end{matrix}$

Signal Σ₅ is sent to LPF 184, as described earlier for LPF 160.

For an LPF T_(n)(x) of order n, the complex output C₅ is approximately:

$\begin{matrix}\begin{matrix}{C_{5} = {T_{n}\left\lbrack \Sigma_{5} \right\rbrack}} \\{= {{A_{1}^{2}{ɛ_{2}\left( {{\cos \left( {{2\zeta_{1}} - \zeta_{2}} \right)} - {{jsin}\left( {{2\zeta_{1}} - \zeta_{2}} \right)}} \right)}} +}} \\{{A_{1}^{2}\begin{bmatrix}{{2ɛ_{- 1}{O\left( \frac{\omega_{c}}{\omega_{D}} \right)}^{n}} + {ɛ_{- 1}{O\left( \frac{\omega_{c}}{3\omega_{D}} \right)}^{n}} + {A_{1}{O\left( \frac{\omega_{c}}{\omega_{D}} \right)}^{n}} +} \\{{2ɛ_{2}{O\left( \frac{\omega_{c}}{2\omega_{D}} \right)}^{n}} + {2ɛ_{3}{O\left( \frac{\omega_{c}}{3\omega_{D}} \right)}^{n}} + {ɛ_{3}{O\left( \frac{\omega_{c}}{\omega_{D}} \right)}^{n}}}\end{bmatrix}}}\end{matrix} & (107)\end{matrix}$

where O(x) denotes a term of the order of x, ω_(c) is the −3 dB angularcutoff frequency, and ω_(D)=dφ/dt.

The term on the right hand sides of Equation (107) with factor A₁ ³ isthe source of the largest errors and accordingly determines thespecifications of n and the minimum ratio for ω_(D)/ω_(c) that can beused when the outputs of LPF 184 is stored in register 186. For a fourthorder LPF, i.e., n=4, and a minimum ratio for ω_(D)/ω_(c)=7, the errorterms on the right hand side of Equation (107) will generate errors thatcorrespond to ≲0.010 nm (3σ). When the stage is moving at a speed suchthat the corresponding Doppler shift frequency ω_(D) /2π is 10 to 100times greater than the bandwidth of the stage servo control system andthe requirement with respect to ω_(D)/ω_(c) is satisfied, the output C₅of LPF 184 is stored in register 186 as C_(5R) under the control ofsignal 161.

Signal Σ₆ is calculated by combining signals Σ⁻¹ and Σ₃ using multiplier172:

$\begin{matrix}\begin{matrix}{\Sigma_{6} = {\Sigma_{- 1}\Sigma_{3}}} \\{= {{A_{1}^{3}{ɛ_{3}\left( {{\cos \left( {{3\zeta_{1}} - \zeta_{3}} \right)} - {{jsin}\left( {{3\zeta_{1}} - \zeta_{3}} \right)}} \right)}} +}} \\{{{A_{1}^{3}{ɛ_{- 1}\left( {{\cos \left( {{4\phi} - \zeta_{- 1} + {3\zeta_{1}}} \right)} - {{jsin}\left( {{4\phi} - \zeta_{- 1} + {3\zeta_{1}}} \right)}} \right)}} +}} \\{{{A_{1}^{4}\left( {{\cos \left( {{2\phi} + {2\zeta_{1}}} \right)} - {{jsin}\left( {{2\phi} + {2\zeta_{1}}} \right)}} \right)} +}} \\{{{A_{1}^{3}{ɛ_{2}\begin{pmatrix}{{3\left( {{\cos \left( {{3\phi} + \zeta_{1} + \zeta_{2}} \right)} - {{jsin}\left( {{3\phi} + \zeta_{1} + \zeta_{2}} \right)}} \right)} +} \\\left( {{\cos \left( {\phi + {3\zeta_{1}} - \zeta_{2}} \right)} - {{jsin}\left( {\phi + {3\zeta_{1}} - \zeta_{2}} \right)}} \right)\end{pmatrix}}} +}} \\{{{3A_{1}^{3}{ɛ_{3}\left( {{\cos \left( {{4\phi} + \zeta_{1} + \zeta_{3}} \right)} - {j\; {\sin \left( {{4\phi} + \zeta_{1} + \zeta} \right)}}} \right)}} +}} \\{{{3A_{1}^{2}{O\left( {ɛ_{i}ɛ_{j}} \right)}} + {\ldots \mspace{11mu}.}}}\end{matrix} & (108)\end{matrix}$

Signal Σ₆ is sent to LPF 174, as described earlier for LPF 160.

For an LPF T_(n)(x) of order n, the complex output C₆ is approximately:

$\begin{matrix}\begin{matrix}{C_{6} = {T_{n}\left\lbrack \Sigma_{6} \right\rbrack}} \\{= {{A_{1}^{3}{ɛ_{3}\left( {{\cos \left( {{3\zeta_{1}} - \zeta_{3}} \right)} - {{jsin}\left( {{3\zeta_{1}} - \zeta_{3}} \right)}} \right)}} +}} \\{{A_{1}^{3}\begin{bmatrix}{{ɛ_{- 1}{O\left( \frac{\omega_{c}}{4\omega_{D}} \right)}^{n}} + {A_{1}{O\left( \frac{\omega_{c}}{2\omega_{D}} \right)}^{n}} +} \\{{3ɛ_{2}{O\left( \frac{\omega_{c}}{3\omega_{D}} \right)}^{n}} + {ɛ_{2}{O\left( \frac{\omega_{c}}{\omega_{D}} \right)}^{n}} + {3ɛ_{3}{O\left( \frac{\omega_{c}}{4\omega_{D}} \right)}^{n}}}\end{bmatrix}}}\end{matrix} & (109)\end{matrix}$

where O(x) denotes a term of the order of x, ω_(c) is the −3 dB angularcutoff frequency, and ω_(D)=dφ/dt.

The term on the right hand sides of Equation (107) with factor A₁ ⁴ isthe source of the largest errors and accordingly determines thespecifications of n and the minimum ratio for ω_(D)/ω_(c) that can beused when the outputs of LPF 174 is stored in register 176. For a fourthorder LPF, i.e., n=4, and a minimum ratio for ω_(D)/ω_(c)=3.5, the errorterms on the right hand side of Equation (109) will generate errors thatcorrespond to ≲0.010 nm (3σ). When the stage is moving at a speed suchthat the corresponding Doppler shift frequency ω_(D)/2π is 10 to 100times greater than the bandwidth of the stage servo control system andthe requirement with respect to ω_(D)/ω_(c) is satisfied, the output C₆of LPF 174 is stored in register 176 as C_(6R) under the control ofsignal 161.

The magnitude squared of signal D(t) is calculated by multiplier 152:

C ₁(t)=D(t)·D(t)*.  (110)

This may be simplified and implemented as:

C ₁(t)=Re(D(t))² +Im(D(t))².  (111)

Alternate methods to calculate a value equivalent to the result ofequation (111) are possible, for example the magnitude output 125 fromCORDIC converter 124 could be squared, or signal Σ₁ could be usedinstead of D(t).

When equation (111) is evaluated using the terms of D(t) from equations(89) to (94), the following result is obtained:

$\begin{matrix}{C_{1} = {A_{1}^{2} + \left\lbrack {ɛ_{- 1}^{2} + ɛ_{0}^{2} + ɛ_{2}^{2} + ɛ_{3}^{2}} \right\rbrack + {2A_{1}ɛ_{- 1}{\cos \left( {{2\phi} + \zeta_{1} - \zeta_{- 1}} \right)}} + {2A_{1}ɛ_{0}{\cos \left( {\phi + \zeta_{1} - \zeta_{0}} \right)}} + {2A_{1}ɛ_{2}{\cos \left( {{- \phi} + \zeta_{1} - \zeta_{2}} \right)}} + {2A_{1}ɛ_{3}{\cos \left( {{{- 2}\phi} + \zeta_{1} - \zeta_{3}} \right)}} + {{O\left( {ɛ_{i}ɛ_{j}} \right)}.}}} & (112)\end{matrix}$

The signal C₁(t) is sent to LPF (Lowpass Filter) 154 as describedearlier for LPF 160.

For an LPF T_(n)(x) of order n, the output C₁ is approximately:

$\begin{matrix}\begin{matrix}{C_{1} = {T_{n}\left\lbrack {C_{1}(t)} \right\rbrack}} \\{= {A_{1}^{2} + \left\lbrack {ɛ_{- 1}^{2} + ɛ_{0}^{2} + ɛ_{2}^{2} + ɛ_{3}^{2}} \right\rbrack + {2A_{1}} + {\begin{bmatrix}{{ɛ_{- 1}{O\left( \frac{\omega_{c}}{2\omega_{D}} \right)}^{n}} + {ɛ_{0}{O\left( \frac{\omega_{c}}{\omega_{D}} \right)}^{n}} +} \\{{ɛ_{2}{O\left( \frac{\omega_{c}}{\omega_{D}} \right)}^{n}} + {ɛ_{3}{O\left( \frac{\omega_{c}}{2\omega_{D}} \right)}^{n}}}\end{bmatrix}.}}}\end{matrix} & (113)\end{matrix}$

The accuracy required for the determination of C₁ is approximately 0.5%in order to limit errors generated in the computation of cyclic errorsignals S_(ψj) to ≲0.010 nm (3σ). Therefore the error terms ε⁻¹ ², ε₀ ²,ε₂ ², and ε₂ ² on the right hand side of Equation (113) are negligible.The terms on the right hand side of Equation (113) of the form

${O\left( \frac{\omega_{c}}{\omega_{D}} \right)}^{n}$

are the sources of the largest Doppler shift frequency dependent errorsand accordingly determine the specifications of n and the minimum ratiofor ω_(D)/ω_(c) that can be used when the output of LPF 154 is held inregister 156, providing signal C_(1R). For a second order LPF, i.e.,n=2, and a minimum ratio for ω_(D)/ω_(c)=3.5, the Doppler shiftfrequency dependent error terms on the right hand side of Equation (113)will generate errors that correspond to ≲0.010 nm (3σ). When the stageis moving at a speed such that the corresponding Doppler shift frequencyω_(D)/2π is 10 to 100 times greater than the bandwidth of the stageservo control system and the requirement with respect to OD/ω_(c) issatisfied, the output C₁ of LPF 154 is stored in register 156 as C_(1R)under the control of signal 161.

The low pass filtering approach to determining values from which errorbasis functions and their coefficients are derived is appropriate whenthe stage is moving at a speed such that the corresponding Doppler shiftfrequency satisfies constraints due to the low pass filter −3 dB cutoffω_(c) and the stage servo control system bandwidth. When the stage isbeing scanned slowly, or is nearly stationary (e.g., due to a change inscan direction), a distribution analysis approach can be used tocalculate and store the values used to generate the compensating signalD_(ψ)(t).

The distribution analysis approach includes performing errorcompensation calculations based on collective properties of adistribution of values derived at least in part from samples of thesignal S(t). The values may represent, for example, samples of amulti-dimensional signal. As a function of time, the multi-dimensionalsignal defines a curve, and the values represent points on the curve.The distribution analysis approach is described in more detail, forexample, in U.S. application Ser. No. 11/462,185, incorporated herein byreference.

Using the coefficients derived using either the low pass filteringapproach or the distribution analysis approach, the processor 204calculates compensating signal D_(ψ)(t) as shown in FIG. 4 e andEquation (114).

$\begin{matrix}{{D\; {\psi (t)}} = {C_{0R} + {\frac{C_{4R}}{C_{1R}}\Sigma_{- 1}} + {\left( {\frac{C_{5R}}{C_{1R}^{2}}*}\; \right)\Sigma_{2}} + {\left( {\frac{C_{6R}}{C_{1R}^{3}}*}\; \right)\Sigma_{3}}}} & (118)\end{matrix}$

This can be expanded using the most significant terms from thepreviously expressed equations (98), (113), (102), (107), and (109) forthe C_(N) terms, and equations (100), (101), and (105) for the Σ_(N)terms to:

$\begin{matrix}{{D\; {\psi (t)}} = {{ɛ_{0}\left( {{\cos \left( \zeta_{0} \right)} - {{jsin}\left( \zeta_{0} \right)}} \right)} + {\frac{A_{1}{ɛ_{- 1}\left( {{\cos \left( {\zeta_{1} + \zeta_{- 1}} \right)} - {{jsin}\left( {\zeta_{1} + \zeta_{- 1}} \right)}} \right)}}{A_{1}^{2}}{A_{1}\left( {{\cos \left( {{- \phi} - \zeta_{1}} \right)} - {{jsin}\left( {{- \phi} - \zeta_{1}} \right)}} \right)}} + {\left( {\frac{A_{1}^{2}{ɛ_{2}\left( {{\cos \left( {{2\zeta_{1}} - \zeta_{2}} \right)} - {{jsin}\left( {{2\zeta_{1}} - \zeta_{2}} \right)}} \right)}}{A_{1}^{4}}*}\; \right){A_{1}^{2}\left( {{\cos \left( {{2\phi} + {2\zeta_{1}}} \right)} - {{jsin}\left( {{2\phi} + {2\zeta_{1}}} \right)}} \right)}} + {\left( {\frac{A_{1}^{3}{ɛ_{3}\left( {{\cos \left( {{3\zeta_{1}} - \zeta_{3}} \right)} - {{jsin}\left( {{3\zeta_{1}} - \zeta_{3}} \right)}} \right)}}{A_{1}^{6}}*}\; \right){A_{1}^{3}\left( {{\cos \left( {{3\phi} + {3\zeta_{1}}} \right)} - {{jsin}\left( {{3\phi} + {3\zeta_{1}}} \right)}} \right)}}}} & (119)\end{matrix}$

and simplified to:

$\begin{matrix}{{D\; {\psi (t)}} = {{ɛ_{0}\left( {{\cos \left( \zeta_{0} \right)} - {{jsin}\left( \zeta_{0} \right)}} \right)} + {ɛ_{- 1}\left( {{\cos \left( {{- \phi} + \zeta_{- 1}} \right)} - {{jsin}\left( {{- \phi} + \zeta_{- 1}} \right)}} \right)} + {ɛ_{2}\left( {{\cos \left( {{2\phi} + \zeta_{2}} \right)} - {{jsin}\left( {{2\phi} + \zeta_{2}} \right)}} \right)} + {{ɛ_{3}\left( {{\cos \left( {{3\phi} + \zeta_{3}} \right)} - {{jsin}\left( {{3\phi} + \zeta_{3}} \right)}} \right)}.}}} & (120)\end{matrix}$

Note that the calculated result in equation (116) is identical toD_(ψ)(t) as described in equations (90) to (94).

Control signal 211 enables the output of D_(ψ)(t) when conditionsindicate that a valid compensation signal is present. Control signal 211is generated by control circuitry (not shown) that may consider thespeed of motion, the signal level, the cyclic error level, the durationof motion, and possible other conditions to determine the validity orusefulness of D_(ψ)(t). It may also be desirable to enable each of thefour cyclic error terms separately, for example if the ε₃ term was foundto be insignificant for a specific implementation, the calculation couldbe disabled to reduce noise contributed by the calculation. When aD_(ψ)(t) term or output is disabled, zero is substituted.

The calculated D_(ψ)(t) is subtracted from signal D(t) by subtractor146, resulting in a measurement signal with substantially reduced cyclicerrors.

Using either the DMS approach or the CMS approach, other techniques canbe incorporated to compensate for some of the sources of cyclic errors.For example, the dominant error term is typically the unshifted cyclicerror component, S_(ψ0), or equivalently D_(ψ0), which stays at constantphase and frequency regardless of stage motion. This term arises fromthe presence of both optical frequencies in either the reference arm orthe measurement arm of the displacement measuring interferometer orboth. This occurs, for example, if the optical frequencies of the lightsource are not perfectly separated into orthogonal linear polarizationstates.

Though the cyclic error compensation techniques are described using thedouble pass plane mirror interferometer by way of example, they can beapplied to any two-frequency, displacement measuring interferometer inwhich the cyclic error term which does not experience Doppler shift isdominant.

Compensation for this unshifted term can be achieved by momentarilyblocking the beam in the reference path of the interferometer and thenin the measurement path of the interferometer at two different times,during, for example, interferometer installation; measuring andrecording the complex amplitude of the measurement signal in thebeam-blocked condition; adding the two different complex amplitudes soobtained to obtain a compensation coefficient representing the unshiftedterm that is present when neither beam is blocked; then using thecomplex coefficient to compensate for the unshifted term duringoperation of the electronic compensation techniques described hereinwith the automatic coefficient updating disabled.

The technique of beam blocking can either be manual or automated using,for example, motorized shutters in the reference and measurement pathsof the interferometer which are controlled by the same computer thatreads the complex amplitudes in the beam-blocked condition; calculatesthe sum; and stores the coefficient for use in the electroniccompensation techniques described herein.

Finally, it is noted that, if desired, the accuracy of the cyclic errorcorrection can further be improved to higher order in the amplitude ofthe cyclic error coefficients by iterating the compensation of the mainsignal S(t). In other words, for each subsequent iteration thecompensated signal for the main signal is used to generate correspondingcyclic error basis functions and determine a higher order correction toeach of the cyclic error coefficients.

Referring now to FIG. 5, a block diagram is shown for an M^(th) orderdigital filter used in the cyclic error compensation described above.The figure is in the “Direct Form I” representation standard to thedigital signal processing community. The input discrete time series isx(n) and the output discrete time series is y(n). The z⁻¹ operationindicates a one-sample delay. A time-domain representation of the filtertakes the form:

y(n)=b ₀ x(n)+b ₁ x(n−1)+b ₂ x(n−2)+ . . . +b _(M) x(n−M)−a ₁ y(n−1)−a ₂y(n−2)− . . . −a _(M) y(n−M)  (121).

The coefficients a_(i) and b_(i) are selected to produce the desiredproperties for the filter. For the case of the Butterworth filter, thea_(i) and b_(i) coefficients are selected to produce the frequencyfiltering given by Equations 28, 29, 43, 44, 55, and 56. Furthermore,other embodiments of the cyclic error compensation may implementdifferent low-pass filtering schemes to yield the coefficients oflow-frequency terms. The Butterworth filter, and other low-pass digitalfilters, are well known in the art. See, for example: Oppenhiem, A. V.,Schafer, R. W., and, J. R. Buck, “Discrete-Time Signal Processing”,Upper Saddle River, N.J.: Prentice Hall, 1999; and Proakis, J. G., and,D. G. Manolakis, “Digital Signal Processing: Principles, Algorithms, andApplications”, New York, N.Y.: MacMillan, 1992.

Depending on the embodiment, the compensation technique described abovecan be implemented in hardware or software, or a combination of both.The technique can be implemented in computer programs using standardprogramming techniques following the method and figures describedherein. Program code is applied to input data to perform the functionsdescribed herein and generate output information. The output informationis applied to one or more output devices such as the servo controlsystem.

Each program may be implemented in a high level procedural or objectoriented programming language to communicate with a computer system, orthe programs can be implemented in assembly or machine language, ifdesired. In any case, the language can be a compiled or interpretedlanguage. Moreover, the program can run on dedicated integrated circuitspreprogrammed for that purpose.

Each such computer program may be stored on a storage medium or device(e.g., ROM or magnetic diskette) readable by a general or specialpurpose programmable computer, for configuring and operating thecomputer when the storage media or device is read by the computer toperform the procedures described herein. The computer program can alsoreside in cache or main memory during program execution. Thecompensation technique can also be implemented as a computer-readablestorage medium, configured with a computer program, where the storagemedium so configured causes a computer to operate in a specific andpredefined manner to perform the functions described herein.

Now referring to FIG. 6, an interferometry system including a highstability plane mirror interferometer (HSPMI) 411 is shown for opticalgenerating the main interference signal. The HSPMI 411 includes apolarization beam-splitter 330, a retroreflector 332, quarter wave phaseretardation plates 334 and 336, and a plane mirror reference object 342.Input beam 422 is a two-component beam. The two components havedifferent frequencies and are orthogonally plane polarized. Thedifferent frequencies can be produced in source 415, for example, bylaser Zeeman splitting, by acousto-optical modulation, or internal tothe laser using birefringent elements or the like. HSPMI 411 splitsinput beam 422 into two components. One component, shown as first andsecond pass measurement beams 322 and 324, reflects from measurementobject 490 twice before exiting HSPMI 411. The other component, shown byfirst and second pass reference beams 328 and 327, reflect fromreference mirror 342 twice before exiting HSPMI 411. The exiting beamcomponents overlap and form output beam 423.

An electrical interference signal 352 is generated by the detection ofoutput beam 423 in detector 420. Detector 420 includes a polarizer tomix the reference and measurement beam components of output beam 423with respect to polarization. Electrical interference signal 352contains a heterodyne interference signal corresponding to maininterference signal S(t).

In further embodiments, the interferometry system may be different thanthat shown in FIG. 6. In particular, the cyclic error compensationtechnique is applicable to other types of interferometers as well. Forexample, the main interference signal S(t) may be produced by aninterferometry system that may include any of single and/or multiplepass interferometers, passive interferometers, dynamic interferometers,and dispersion interferometers. Furthermore, the interferometry systemmay monitor one or more degrees of freedom, each of which may produce acorresponding main interference signal S(t), which may be compensatedfor cyclic errors as disclosed herein. Furthermore, the degree(s) offreedom monitored by the interferometry system may include any ofchanges in distance to a measurement object, changes in relativedistance between two measurement objects, changes in the angularorientation of a measurement object, and changes in the direction of theinput beam.

Examples of dynamic interferometers are described in U.S. patentapplication Ser. No. 10/226,591 filed Aug. 23, 2002 and entitled“DYNAMIC INTERFEROMETER CONTROLLING DIRECTION OF INPUT BEAM” by Henry A.Hill. Examples of passive zero shear interferometers are described inU.S. patent application Ser. No. 10/207,314, entitled “PASSIVE ZEROSHEAR INTERFEROMETERS,” filed Jul. 29, 2002, by Henry A. Hill. Examplesof angular displacement interferometers are described in: U.S. patentapplication Ser. No. 10/226,591 entitled “DYNAMIC INTERFEROMETERCONTROLLING DIRECTION OF INPUT BEAM,” filed Aug. 23, 2002; U.S.Provisional Application 60/314,345 filed Aug. 22, 2001 and entitled“PASSIVE ZERO SHEAR INTERFEROMETERS USING ANGLE SENSITIVEBEAM-SPLITTERS,” both by Henry A. Hill, and U.S. patent application Ser.No. 10/272,034 entitled “INTERFEROMETERS FOR MEASURING CHANGES INOPTICAL BEAM DIRECTION” and filed Oct. 15, 2002 by Henry A. Hill andJustin Kreuzer. Alternatively, or additionally, interferometry systemsmay include one or more differential angular displacementinterferometers, examples of which are also described in U.S. patentapplication Ser. No. 10/272,034. Examples of interferometry systems formeasuring more than one degree of freedom and for reducing beam shearare described in U.S. patent application Ser. No. 10/352,616 filed Jan.28, 2003 and entitled “MULTIPLE-PASS INTERFEROMETRY” by Henry A. Hilland U.S. patent application Ser. No. 10/351,708 filed Jan. 27, 2003 andentitled “MULTI-AXIS INTERFEROMETER” by Henry A. Hill. Other forms ofmultiple pass interferometers are described in an article entitled“Differential interferometer arrangements for distance and anglemeasurements: Principles, advantages and applications” by C. Zanoni, VDIBerichte Nr. 749, 93-106 (1989). Examples of two-wavelength dispersioninterferometers are described in U.S. Pat. No. 6,219,144 B1 entitled“APPARATUS AND METHOD FOR MEASURING THE REFRACTIVE INDEX AND OPTICALPATH LENGTH EFFECTS OF AIR USING MULTIPLE-PASS INTERFEROMETRY” by HenryA. Hill, Peter de Groot, and Frank C. Demarest and U.S. Pat. No.6,327,039 B1 by Peter de Groot, Henry A. Hill, and Frank C. Demarest.

Because of the cyclic error compensation, the interferometry systemsdescribed herein provide highly accurate measurements. Such systems canbe especially useful in lithography applications used in fabricatinglarge scale integrated circuits such as computer chips and the like.Lithography is the key technology driver for the semiconductormanufacturing industry. Overlay improvement is one of the five mostdifficult challenges down to and below 100 nm line widths (designrules), see, for example, the Semiconductor Industry Roadmap, p. 82(1997).

Overlay depends directly on the performance, i.e., accuracy andprecision, of the distance measuring interferometers used to positionthe wafer and reticle (or mask) stages. Since a lithography tool mayproduce $50-100M/year of product, the economic value from improvedperformance distance measuring interferometers is substantial. Each 1%increase in yield of the lithography tool results in approximately$1M/year economic benefit to the integrated circuit manufacturer andsubstantial competitive advantage to the lithography tool vendor.

The function of a lithography tool is to direct spatially patternedradiation onto a photoresist-coated wafer. The process involvesdetermining which location of the wafer is to receive the radiation(alignment) and applying the radiation to the photoresist at thatlocation (exposure).

To properly position the wafer, the wafer includes alignment marks onthe wafer that can be measured by dedicated sensors. The measuredpositions of the alignment marks define the location of the wafer withinthe tool. This information, along with a specification of the desiredpatterning of the wafer surface, guides the alignment of the waferrelative to the spatially patterned radiation. Based on suchinformation, a translatable stage supporting the photoresist-coatedwafer moves the wafer such that the radiation will expose the correctlocation of the wafer.

During exposure, a radiation source illuminates a patterned reticle,which scatters the radiation to produce the spatially patternedradiation. The reticle is also referred to as a mask, and these termsare used interchangeably below. In the case of reduction lithography, areduction lens collects the scattered radiation and forms a reducedimage of the reticle pattern. Alternatively, in the case of proximityprinting, the scattered radiation propagates a small distance (typicallyon the order of microns) before contacting the wafer to produce a 1:1image of the reticle pattern. The radiation initiates photo-chemicalprocesses in the resist that convert the radiation pattern into a latentimage within the resist.

Interferometry systems are important components of the positioningmechanisms that control the position of the wafer and reticle, andregister the reticle image on the wafer. If such interferometry systemsinclude the features described above, the accuracy of distances measuredby the systems increases as cyclic error contributions to the distancemeasurement are minimized.

In general, the lithography system, also referred to as an exposuresystem, typically includes an illumination system and a waferpositioning system. The illumination system includes a radiation sourcefor providing radiation such as ultraviolet, visible, x-ray, electron,or ion radiation, and a reticle or mask for imparting the pattern to theradiation, thereby generating the spatially patterned radiation. Inaddition, for the case of reduction lithography, the illumination systemcan include a lens assembly for imaging the spatially patternedradiation onto the wafer. The imaged radiation exposes resist coatedonto the wafer. The illumination system also includes a mask stage forsupporting the mask and a positioning system for adjusting the positionof the mask stage relative to the radiation directed through the mask.The wafer positioning system includes a wafer stage for supporting thewafer and a positioning system for adjusting the position of the waferstage relative to the imaged radiation. Fabrication of integratedcircuits can include multiple exposing steps. For a general reference onlithography, see, for example, J. R. Sheats and B. W. Smith, inMicrolithography: Science and Technology (Marcel Dekker, Inc., New York,1998), the contents of which is incorporated herein by reference.

Interferometry systems described above can be used to precisely measurethe positions of each of the wafer stage and mask stage relative toother components of the exposure system, such as the lens assembly,radiation source, or support structure. In such cases, theinterferometry system can be attached to a stationary structure and themeasurement object attached to a movable element such as one of the maskand wafer stages. Alternatively, the situation can be reversed, with theinterferometry system attached to a movable object and the measurementobject attached to a stationary object.

More generally, such interferometry systems can be used to measure theposition of any one component of the exposure system relative to anyother component of the exposure system, in which the interferometrysystem is attached to, or supported by, one of the components and themeasurement object is attached, or is supported by the other of thecomponents.

An example of a lithography scanner 1100 using an interferometry system1126 is shown in FIG. 7. The interferometry system is used to preciselymeasure the position of a wafer (not shown) within an exposure system.Here, stage 1122 is used to position and support the wafer relative toan exposure station. Scanner 1100 includes a frame 1102, which carriesother support structures and various components carried on thosestructures. An exposure base 1104 has mounted on top of it a lenshousing 1106 atop of which is mounted a reticle or mask stage 1116,which is used to support a reticle or mask. A positioning system forpositioning the mask relative to the exposure station is indicatedschematically by element 1117. Positioning system 1117 can include,e.g., piezoelectric transducer elements and corresponding controlelectronics. Although, it is not included in this described embodiment,one or more of the interferometry systems described above can also beused to precisely measure the position of the mask stage as well asother moveable elements whose position must be accurately monitored inprocesses for fabricating lithographic structures (see supra Sheats andSmith Microlithography: Science and Technology).

Suspended below exposure base 1104 is a support base 1113 that carrieswafer stage 1122. Stage 1122 includes a plane mirror 1128 for reflectinga measurement beam 1154 directed to the stage by interferometry system1126. A positioning system for positioning stage 1122 relative tointerferometry system 1126 is indicated schematically by element 1119.Positioning system 1119 can include, e.g., piezoelectric transducerelements and corresponding control electronics. The measurement beamreflects back to the interferometry system, which is mounted on exposurebase 1104. The interferometry system can be any of the embodimentsdescribed previously.

During operation, a radiation beam 1110, e.g., an ultraviolet (UV) beamfrom a UV laser (not shown), passes through a beam shaping opticsassembly 1112 and travels downward after reflecting from mirror 1114.Thereafter, the radiation beam passes through a mask (not shown) carriedby mask stage 1116. The mask (not shown) is imaged onto a wafer (notshown) on wafer stage 1122 via a lens assembly 1108 carried in a lenshousing 1106. Base 1104 and the various components supported by it areisolated from environmental vibrations by a damping system depicted byspring 1120.

In other embodiments of the lithographic scanner, one or more of theinterferometry systems described previously can be used to measuredistance along multiple axes and angles associated for example with, butnot limited to, the wafer and reticle (or mask) stages. Also, ratherthan a UV laser beam, other beams can be used to expose the waferincluding, e.g., x-ray beams, electron beams, ion beams, and visibleoptical beams.

In some embodiments, the lithographic scanner can include what is knownin the art as a column reference. In such embodiments, theinterferometry system 1126 directs the reference beam (not shown) alongan external reference path that contacts a reference mirror (not shown)mounted on some structure that directs the radiation beam, e.g., lenshousing 1106. The reference mirror reflects the reference beam back tothe interferometry system. The interference signal produce byinterferometry system 1126 when combining measurement beam 1154reflected from stage 1122 and the reference beam reflected from areference mirror mounted on the lens housing 1106 indicates changes inthe position of the stage relative to the radiation beam. Furthermore,in other embodiments the interferometry system 1126 can be positioned tomeasure changes in the position of reticle (or mask) stage 1116 or othermovable components of the scanner system. Finally, the interferometrysystems can be used in a similar fashion with lithography systemsinvolving steppers, in addition to, or rather than, scanners.

As is well known in the art, lithography is a critical part ofmanufacturing methods for making semiconducting devices. For example,U.S. Pat. No. 5,483,343 outlines steps for such manufacturing methods.These steps are described below with reference to FIGS. 8 a and 8 b.FIG. 8 a is a flow chart of the sequence of manufacturing asemiconductor device such as a semiconductor chip (e.g., IC or LSI), aliquid crystal panel or a CCD. Step 1151 is a design process fordesigning the circuit of a semiconductor device. Step 1152 is a processfor manufacturing a mask on the basis of the circuit pattern design.Step 1153 is a process for manufacturing a wafer by using a materialsuch as silicon.

Step 1154 is a wafer process which is called a pre-process wherein, byusing the so prepared mask and wafer, circuits are formed on the waferthrough lithography. To form circuits on the wafer that correspond withsufficient spatial resolution those patterns on the mask,interferometric positioning of the lithography tool relative the waferis necessary. The interferometry methods and systems described hereincan be especially useful to improve the effectiveness of the lithographyused in the wafer process.

Step 1155 is an assembling step, which is called a post-process whereinthe wafer processed by step 1154 is formed into semiconductor chips.This step includes assembling (dicing and bonding) and packaging (chipsealing). Step 1156 is an inspection step wherein operability check,durability check and so on of the semiconductor devices produced by step1155 are carried out. With these processes, semiconductor devices arefinished and they are shipped (step 1157).

FIG. 8 b is a flow chart showing details of the wafer process. Step 1161is an oxidation process for oxidizing the surface of a wafer. Step 1162is a CVD process for forming an insulating film on the wafer surface.Step 1163 is an electrode forming process for forming electrodes on thewafer by vapor deposition. Step 1164 is an ion implanting process forimplanting ions to the wafer. Step 1165 is a resist process for applyinga resist (photosensitive material) to the wafer. Step 1166 is anexposure process for printing, by exposure (i.e., lithography), thecircuit pattern of the mask on the wafer through the exposure apparatusdescribed above. Once again, as described above, the use of theinterferometry systems and methods described herein improve the accuracyand resolution of such lithography steps.

Step 1167 is a developing process for developing the exposed wafer. Step1168 is an etching process for removing portions other than thedeveloped resist image. Step 1169 is a resist separation process forseparating the resist material remaining on the wafer after beingsubjected to the etching process. By repeating these processes, circuitpatterns are formed and superimposed on the wafer.

The interferometry systems described above can also be used in otherapplications in which the relative position of an object needs to bemeasured precisely. For example, in applications in which a write beamsuch as a laser, x-ray, ion, or electron beam, marks a pattern onto asubstrate as either the substrate or beam moves, the interferometrysystems can be used to measure the relative movement between thesubstrate and write beam.

As an example, a schematic of a beam writing system 1200 is shown inFIG. 9. A source 1210 generates a write beam 1212, and a beam focusingassembly 1214 directs the radiation beam to a substrate 1216 supportedby a movable stage 1218. To determine the relative position of thestage, an interferometry system 1220 directs a reference beam 1222 to amirror 1224 mounted on beam focusing assembly 1214 and a measurementbeam 1226 to a mirror 1228 mounted on stage 1218. Since the referencebeam contacts a mirror mounted on the beam focusing assembly, the beamwriting system is an example of a system that uses a column reference.Interferometry system 1220 can be any of the interferometry systemsdescribed previously. Changes in the position measured by theinterferometry system correspond to changes in the relative position ofwrite beam 1212 on substrate 1216. Interferometry system 1220 sends ameasurement signal 1232 to controller 1230 that is indicative of therelative position of write beam 1212 on substrate 1216. Controller 1230sends an output signal 1234 to a base 1236 that supports and positionsstage 1218. In addition, controller 1230 sends a signal 1238 to source1210 to vary the intensity of, or block, write beam 1212 so that thewrite beam contacts the substrate with an intensity sufficient to causephotophysical or photochemical change only at selected positions of thesubstrate.

Furthermore, in some embodiments, controller 1230 can cause beamfocusing assembly 1214 to scan the write beam over a region of thesubstrate, e.g., using signal 1244. As a result, controller 1230 directsthe other components of the system to pattern the substrate. Thepatterning is typically based on an electronic design pattern stored inthe controller. In some applications the write beam patterns a resistcoated on the substrate and in other applications the write beamdirectly patterns, e.g., etches, the substrate.

An important application of such a system is the fabrication of masksand reticles used in the lithography methods described previously. Forexample, to fabricate a lithography mask an electron beam can be used topattern a chromium-coated glass substrate. In such cases where the writebeam is an electron beam, the beam writing system encloses the electronbeam path in a vacuum. Also, in cases where the write beam is, e.g., anelectron or ion beam, the beam focusing assembly includes electric fieldgenerators such as quadrapole lenses for focusing and directing thecharged particles onto the substrate under vacuum. In other cases wherethe write beam is a radiation beam, e.g., x-ray, UV, or visibleradiation, the beam focusing assembly includes corresponding optics andfor focusing and directing the radiation to the substrate.

A number of embodiments of the invention have been described.Nevertheless, it will be understood that various modifications may bemade without departing from the spirit and scope of the invention.Accordingly, other embodiments are within the scope of the followingclaims.

1. A method comprising: directing a first portion of a beam including afirst frequency component along a first path; frequency shifting asecond portion of the beam to generate a shifted beam that includes asecond frequency component different from the first frequency componentand one or more spurious frequency components different from the firstfrequency component; directing at least a portion of the shifted beamalong a second path different from the first path; measuring aninterference signal S(t) from interference between the beam portionsdirected along the different paths, wherein the signal S(t) isindicative of changes in an optical path difference n{tilde over (L)}(t)between the paths, where n is an average refractive index along thepaths, {tilde over (L)}(t) is a total physical path difference betweenthe paths, and t is time; and providing an error signal to reduce errorsin an estimate of {tilde over (L)}(t) that are caused by at least one ofthe spurious frequency components of the shifted beam, the error signalbeing derived at least in part based on the signal S(t).
 2. The methodof claim 1, further comprising providing the estimate of {tilde over(L)}(t) based at least in part on subtracting the error signal from thesignal S(t).
 3. The method of claim 1, wherein frequency shifting thesecond beam portion comprises modulating the phase of the second beamportion to generate the shifted beam.
 4. The method of claim 3, whereinthe phase modulation comprises electro-optic phase modulation.
 5. Themethod of claim 3 wherein the phase modulation comprises Serrodynemodulation.
 6. The method of claim 5, wherein the errors are dependenton a non-zero fall time of a ramp associated with the Serrodynemodulation.
 7. The method of claim 6, wherein the errors are furtherdependent on an amplitude of the ramp.
 8. The method of claim 3, furthercomprising providing the estimate of {tilde over (L)}(t) based at leastin part on adjusting the phase modulation of the second beam portionusing the error signal.
 9. The method of claim 8, wherein the phasemodulation is adjusted using the phase of the error signal.
 10. Themethod of claim 9, wherein an amplitude of the phase modulation isadjusted using the phase of the error signal.
 11. The method of claim 1,wherein at least one of the beam portions is directed to reflect from amovable measurement object before producing the interference signal S(t)to generate a Doppler shift indicative of movement of the measurementobject.
 12. The method of claim 11, further comprising providing theestimate of {tilde over (L)}(t) based at least in part on a sinusoidalcomponent of the signal S(t) that has a time-varying argumentcorresponding to a sum of a term proportional to a difference betweenpeak frequencies of the first frequency component and the secondfrequency component and a term proportional to a Doppler shiftfrequency.
 13. The method of claim 12, wherein the error signal includesa sinusoidal component that has a time-varying argument corresponding toa difference between the term proportional to the difference between thepeak frequencies of the first frequency component and the secondfrequency component and the term proportional to the Doppler shiftfrequency.
 14. The method of claim 11, wherein the difference betweenthe peak frequencies of the first frequency component and the secondfrequency component is less than about 500 kHz.
 15. The method of claim14, wherein the Doppler shift frequency is less than about 50 kHz. 16.The method of claim 1, wherein one of the paths is associated with aposition of a reference object and the other path is associated with aposition of a moveable measurement object.
 17. The method of claim 16,wherein the position of the moveable object is controlled by a servosystem.
 18. The method of claim 17, wherein the servo system controlsthe position of the measurement object based on the signal S(t) and theerror signal.
 19. The method of claim 1, wherein providing the errorsignal to reduce the errors caused by at least one spurious frequencycomponent in the shifted beam comprises providing one or morecoefficients representative of one or more errors that cause the signalS(t) to deviate from an ideal expression of the form A₁cos(ω_(R)t+φ(t)+ζ₁), where A₁ and ζ₁ are constants, ω_(R) is an angularfrequency difference between the first frequency component and thesecond frequency component, and φ(t)=nk{tilde over (L)}(t), with k=2π/λand λ equal to a wavelength for the beam.
 20. The method of claim 19,wherein the deviation can be expressed as${\sum\limits_{m,p}{A_{m,p}{\cos \left( {{\omega_{R}t} + {\frac{m}{p}{\phi (t)}} + \zeta_{m,p}} \right)}}},$where p=1, 2, 3 . . . , and m is any integer not equal top, and wherethe provided coefficients comprise information corresponding to at leastsome of A_(m,p) and ζ_(m,p).
 21. The method of claim 19, wherein thecoefficients are derived at least in part based on a plurality ofsamples of the signal S(t).
 22. The method of claim 21, wherein theerror signal is generated from the coefficients and one or more errorbasis functions derived at least in part from a plurality of samples ofthe signal S(t).
 23. The method of claim 22, wherein each of the errorbasis functions is derived at least in part from a linear combination ofsamples of the signal S(t).
 24. The method of claim 22, wherein each ofthe error basis functions corresponds to a function that includes one ormore leading sinusoidal terms having a time-varying argument thatcorresponds to a time-varying argument of an error term that representsa portion of the deviation of S(t) from the ideal expression.
 25. Themethod of claim 1, wherein reducing errors in the estimate of {tildeover (L)}(t) comprises deriving the estimate of {tilde over (L)}(t) froma difference between the error signal and a discrete Fourier transformof samples of S(t).
 26. A lithography method for use in fabricatingintegrated circuits on a wafer, the method comprising: supporting thewafer on a moveable stage; imaging spatially patterned radiation ontothe wafer; adjusting the position of the stage; and monitoring theposition of the stage using an interferometry system, wherein monitoringthe position of the stage comprises reducing errors in an estimate of aphysical path difference associated with a position of a measurementobject associated with the stage using the method of claim
 1. 27. Amethod for fabricating integrated circuits, the method comprising:applying a resist to a wafer; forming a pattern of a mask in the resistby exposing the wafer to radiation using the lithography method of claim26; and producing an integrated circuit from the wafer.
 28. Alithography method for use in the fabrication of integrated circuits,the method comprising: directing input radiation through a mask toproduce spatially patterned radiation; positioning the mask relative tothe input radiation; monitoring the position of the mask relative to theinput radiation using an interferometry system, wherein monitoring theposition of the mask comprises reducing errors in an estimate of aphysical path difference associated with the position of the mask usingthe method of claim 1; and imaging the spatially patterned radiationonto a wafer.
 29. A method for fabricating integrated circuits, themethod comprising: applying a resist to a wafer; forming a pattern of amask in the resist by exposing the wafer to radiation using thelithography method of claim 28; and producing an integrated circuit fromthe wafer.
 30. A lithography method for fabricating integrated circuitson a wafer, the method comprising: positioning a first component of alithography system relative to a second component of a lithographysystem to expose the wafer to spatially patterned radiation; andmonitoring the position of the first component relative to the secondcomponent using an interferometry system, wherein monitoring theposition of the first component comprises reducing errors in an estimateof a physical path difference associated with a position of ameasurement object associated with the first component using the methodof claim
 1. 31. The method of claim 30, wherein the first componentcomprises projection optics, and the second component comprises a stagefor supporting the wafer.
 32. A method for fabricating integratedcircuits, the method comprising: applying a resist to a wafer; forming apattern of a mask in the resist by exposing the wafer to radiation usingthe lithography method of claim 30; and producing an integrated circuitfrom the wafer.
 33. A method for fabricating a lithography mask, themethod comprising: directing a write beam to a substrate to pattern thesubstrate; positioning the substrate relative to the write beam; andmonitoring the position of the substrate relative to the write beamusing an interferometry system, wherein monitoring the position of thesubstrate comprises reducing errors in an estimate of a physical pathdifference associated with a position of a measurement object associatedwith the substrate using the method of claim
 1. 34. An apparatuscomprising a computer readable medium which during operation causes aprocessor to perform the method of claim
 1. 35. An apparatus comprising:an interferometry system, which during operation directs a first portionof a beam including a first frequency component along a first path;frequency shifts a second portion of the beam to generate a shifted beamthat includes a second frequency component different from the firstfrequency component and one or more spurious frequency componentsdifferent from the first frequency component; and directs at least aportion of the shifted beam along a second path different from the firstpath; a detector that measures an interference signal S(t) frominterference between the beam portions directed along the differentpaths, wherein the signal S(t) is indicative of changes in an opticalpath difference n{tilde over (L)}(t) between the paths, where n is anaverage refractive index along the paths, {tilde over (L)}(t) is a totalphysical path difference between the paths, and t is time; and anelectronic processor, which during operation receives the interferencesignal S(t) from the detector and provides an error signal to reduceerrors in an estimate of {tilde over (L)}(t) that are caused by at leastone of the spurious frequency components of the shifted beam, the errorsignal being derived at least in part based on the signal S(t).
 36. Theapparatus of claim 35, wherein the second beam portion is frequencyshifted in a phase modulator that modulates the phase of the second beamportion to generate the shifted beam.
 37. The apparatus of claim 36,wherein the phase modulator comprises an electro-optic phase modulator.38. A lithography system for use in fabricating integrated circuits on awafer, the system comprising: a stage for supporting the wafer; anillumination system for imaging spatially patterned radiation onto thewafer; a positioning system for adjusting the position of the stagerelative to the imaged radiation; and the apparatus of claim 35 formonitoring the position of the wafer relative to the imaged radiation.39. A method for fabricating integrated circuits, the method comprising:applying a resist to a wafer; forming a pattern of a mask in the resistby exposing the wafer to radiation using the lithography system of claim38; and producing an integrated circuit from the wafer.
 40. Alithography system for use in fabricating integrated circuits on awafer, the system comprising: a stage for supporting the wafer; and anillumination system including a radiation source, a mask, a positioningsystem, a lens assembly, and the apparatus of claim 35, wherein duringoperation the source directs radiation through the mask to producespatially patterned radiation, the positioning system adjusts theposition of the mask relative to the radiation from the source, the lensassembly images the spatially patterned radiation onto the wafer, andthe apparatus monitors the position of the mask relative to theradiation from the source.
 41. A method for fabricating integratedcircuits, the method comprising: applying a resist to a wafer; forming apattern of a mask in the resist by exposing the wafer to radiation usingthe lithography system of claim 40; and producing an integrated circuitfrom the wafer.
 42. A beam writing system for use in fabricating alithography mask, the system comprising: a source providing a write beamto pattern a substrate; a stage supporting the substrate; a beamdirecting assembly for delivering the write beam to the substrate; apositioning system for positioning the stage and beam directing assemblyrelative one another; and the apparatus of claim 35 for monitoring theposition of the stage relative to the beam directing assembly.
 43. Amethod for fabricating a lithography mask comprising: directing a beamto a substrate using the beam writing system of claim 42; varying theintensity or the position of the beam at the substrate to form a patternin the substrate; and forming the lithography mask from the patternedsubstrate.